Title:
WAVEGUIDE ANTENNA
United States Patent 3848256
Abstract:
This invention relates to a frequency scanning antenna based on the evanescent mode waveguide filter, which provides the necessary slow wave structure for such an antenna. The filter is provided with suitable slot radiators in the wall of the waveguide giving a radiation beam whose angle varies rapidly with frequency.
US Patent References:
Excitation of a surface wave on a thin plasma sheath surrounding a missile
Hoffman - September 1965 - 3208068
Excitation of a surface wave on a thin plasma sheath surrounding a missile
Hoffman - September 1965 - 3208068
Inventors:
Craven, George Frederick (Harlow, Essex, EN)
Hockham, George Alfred (Takeley, Essex, EN)
Hockham, George Alfred (Takeley, Essex, EN)
Application Number:
05/423344
Publication Date:
11/12/1974
Filing Date:
12/10/1973
View Patent Images:
Export Citation:
Assignee:
International Standard Electric Corporation (New York, NY)
Primary Class:
Other Classes:
343/771, 342/375
International Classes:
H01Q3/22; H01Q21/00; H01Q13/10
Field of Search:
343/768,770,771,854
Primary Examiner:
Lieberman, Eli
Attorney, Agent or Firm:
O'halloran Jr., John Lombardi Menotti Goldberg Edward T. J.
Claims:
What is claimed is
1. A frequency scanning waveguide antenna comprising:
2. A frequency scanning waveguide antenna according to claim 1, wherein said elements are tuning screws.
3. A frequency scanning waveguide antenna according to claim 1, wherein said elements are transverse dielectric slices.
4. A frequency scanning waveguide antenna according to claim 1, wherein said waveguide is rectangular.
5. A frequency scanning waveguide antenna according to claim 4, wherein said slots are located on alternate sides of the longitudinal axis of said waveguide.
6. A frequency scanning waveguide antenna according to claim 4, wherein said slots are located on the same side of the longitudinal axis of said waveguide.
7. A frequency scanning waveguide antenna according to claim 1, further comprising a plurality of lossy capacitive elements for terminating said antenna.
1. A frequency scanning waveguide antenna comprising:
2. A frequency scanning waveguide antenna according to claim 1, wherein said elements are tuning screws.
3. A frequency scanning waveguide antenna according to claim 1, wherein said elements are transverse dielectric slices.
4. A frequency scanning waveguide antenna according to claim 1, wherein said waveguide is rectangular.
5. A frequency scanning waveguide antenna according to claim 4, wherein said slots are located on alternate sides of the longitudinal axis of said waveguide.
6. A frequency scanning waveguide antenna according to claim 4, wherein said slots are located on the same side of the longitudinal axis of said waveguide.
7. A frequency scanning waveguide antenna according to claim 1, further comprising a plurality of lossy capacitive elements for terminating said antenna.
Description:
BACKGROUND OF THE INVENTION
This invention relates to waveguide antennas, and more particularly to frequency scanning waveguide antennas.
Frequency scan phased array antennas are directional antennas in which the angle of the radiated beam is a function of frequency. This is achieved by controlling the relative phase of excitation of individual elements in the array according to a predetermined phase-frequency characteristic. In order that the desired incremental phase change can be obtained with only a moderate frequency scan, it is necessary to employ a large total phase shift (a long electrical path) in each feeder section. Since the free space distance between adjacent elements of the array is only of the order of one half wavelength it becomes necessary greatly to increase the electrical length of the feeder sections which excite adjacent elements in the array.
One way of doing this is to employ a tortuous feeder path, i.e., one with many bends (a serpentine). An alternative, and more sophisticated, method is to provide a feeder with a slow wave structure such as that occurring in a surface wave.
Existing serpentine structures are extremely complex and difficult to make, and existing surface wave structures are also difficult to make.
SUMMARY OF THE INVENTION
It is an object of the present invention to obtain a suitable slow wave structure in a simple and convenient way.
According to a broad aspect of the invention there is provided a frequency scanning waveguide antenna comprising a waveguide having a longitudinal axis, said waveguide being evanescent over the operating frequency band of said antenna; a plurality of capacitive elements within said waveguide each said element spaced apart from adjacent elements by a predetermined distance l and adjusted to provide a conjugate match condition for evanescent mode resonance; and a plurality of longitudinal radiating slots in the wall of said waveguide, each of said slots located midway between adjacent ones of said capacitive elements.
The invention will be better understood from the following description in conjunction with the accompanying drawings, in which:
BRIEF DESCRIPTION OF THE DRAWINGS
FIGS. 1 and 2 are schematic side and plan views respectively of a frequency scan waveguide antenna element with radiating slots;
FIG. 3 shows beam positions for the antenna of FIGS. 1 and 2;
FIG. 4 shows beam positions for an antenna having a different period of radiating slots;
FIG. 5 shows the measured beam angle as a function of frequency for the antenna of FIGS. 1 and 2;
FIG. 6 shows the measured polar diagram of the antenna for different frequencies; and
FIG. 7 shows a termination arrangement for the antenna.
DESCRIPTION OF THE PREFERRED EMBODIMENT
The waveguide antenna shown in FIGS. 1 and 2 comprises a multisection evanescent mode bandpass filter, basically as described in U.S. Pat. No. 3,621,483 constructed in rectangular cross-section waveguide 1 which is evanescent over the operating frequency band of the antenna. The waveguide contains capacitive elements, tuning screws 2, spaced apart by a distance l and adjusted to give the necessary conjugate match condition to achieve evanescent mode resonance for energy transfer along the waveguide over the required passband. Other forms of capacitive element may be used, for example, transverse slices of dielectric material.
Radiating longitudinal slots 3 are provided, as by milling, on alternate sides of the longitudinal center line of the waveguide either in the broad wall of the waveguide, as shown, or in the narrow wall. The transverse center line of each slot is midway between adjacent resonators (the screws 2).
When tuned, there is obtained a slow wave structure using a periodic network of evanescent mode resonators.
Image parameter theory is applicable to such networks analyzed on an impedance basis. The bandwidth of the network depends on the length, l, between resonators (the screws 2) and the propagation constant y.
The equations giving the image impedance, Z i , and the image transfer constant, θ, are given by
Z i = Z o √ tanh yl/2 [ Z o B 1 sinh yl/2 - cosh yl/2/sinh yl/2 - Z o B 1 cosh yl/2]
and
cosh θ = 2[ cosh 2 yl/2 - Z o B 1 sinh yl/2 cosh yl/2] - 1
The network will transmit energy freely over the frequency range in which its image impedance is real. The limits of this range occur at
cosh θ=±1.
Thus, provided -1 < cosh θ < 1, energy will be freely transmitted. For cosh θ to have values in this range, θ must be imaginary, i.e.,
θ=jβ
Thus, since cosh jβ=cos β=±1, at the band limit the limiting values for β are 0 and π, with a continuous range of variation in between.
The importance of the image transfer constant in this analysis lies in the fact that it gives the relationship between the voltages at the input and output of each section of the filter, (i.e., fields in the guide at a and b).
It will, therefore, be clear that as the frequency is varied over the transmission band of the filter the relative phase difference between planes a and b will vary smoothly between 0° and 180°, being approximately 90° at the band center. In a periodic structure such as the one shown, these phase differentials will be repeated at corresponding planes a 1 , b 1 ; a 2 , b 2 ; etc., throughout the structure.
To obtain a corresponding variation over a narrow band in a conventional straight propagating waveguide would necessitate a guide length 20-100 times the value required in evanescent waveguide functioning as described above.
The polar diagram of the antenna is determined by the relative phase and amplitude of the fields in the slots 3. If φ is the phase shift per slot period L then the beam angles Ψ p with respect to the normal of the array (X-axis) are given by the following equation for the structure shown in FIGS. 1 and 2.
sin Ψ p ± = τo/2L (φ/u ± 2p-1)
p = 1, 2, 3.
A real beam exists in direction Ψ p for each │ sin Ψ p │ ≤1. For L/τo = 1/√2 the beam positions are shown in FIG. 3.
Thus for this example at the lower cut-off frequency there exist two beams in space, Ψ 1 - at -45° and Ψ 1 + at 45°. As the frequency is increased to the upper cut-off frequency the beam Ψ 1 - moves to Ψ° while Ψ 1 + moves to +90° and then into imaginary space. There exists a region of about 25° (shown shaded) where Ψ 1 - is the only beam in space and for this slot period represents the useful range.
FIG. 4 shows the scan range over the passband for the example L/τo = 1/2.
At the lower cut-off frequency of the pass band there exists two beams in directions Ψ = ± 90°. As the frequency is increased Ψ 1 + moves beyond cut-off (into imaginary space), and Ψ 1 - increases and reaches Ψ° at the upper cut-off frequency. Thus in this example Ψ 1 - is the only beam in real space.
An antenna was made consisting of 19 periods with L/τo = 1/√2, and the measured beam angle of this antenna is shown as a function of frequency in FIG. 5. This linear characteristic has a slope equal to 27.6°/100 MHz at 5.7 GHz, i.e., 27.6° for 1.75 percent change of frequency.
The polar diagram is readily obtained once a knowledge of the field in the slots is known. For a structure having radiating slots each located on the same side of the longitudinal center line of the waveguide, with a period L, the beam angles are given by
sin Ψ = τo/2L (φ ± 2p)
p = 0, 1, 2, 3.
Typical measured polar diagrams are shown in FIG. 6, showing the change in beam angle with successive increases of frequency, each of 0.25 GHz, from 5.25 to 5.75 GHz, diagrams A, B and C respectively.
It will be noticed that in these polar diagrams there is an oppositely angled low-power beam. This arises because the antenna had resonators which were each of low loss capacitive elements (tuning screws), and the spurious beams resulted from energy being reflected from the far end of the antenna.
This unwanted effect is substantially eliminated by an appropriate load termination of the antenna. This is indicated in FIG. 7 by forming the final resonators with lossy capacitive elements, e.g., lossy tuning screws 2A. Typically the last three, or possibly five, resonators are so constituted, so that residual energy is progressively absorbed and not reflected. There are no radiating slots in this terminating portion of the antenna. The waveguide end is normally closed by a metal plate 4.
The antenna as above described possesses the following advantages:
a. uses commercially available waveguide,
b. the tuning elements introduced into the antenna compensate for the waveguide tolerances,
c. no complex machining process is necessary; only milling the slots,
d. is significantly lower in weight than the serpentine or surface wave antenna, and
e. uses a waveguide of width less than τo/2 which permits stacking of linear arrays for wide angle azimuth scanning.
It is to be understood that the foregoing description of specific examples of this invention is made by way of example only and is not to be considered as a limitation on its scope.
This invention relates to waveguide antennas, and more particularly to frequency scanning waveguide antennas.
Frequency scan phased array antennas are directional antennas in which the angle of the radiated beam is a function of frequency. This is achieved by controlling the relative phase of excitation of individual elements in the array according to a predetermined phase-frequency characteristic. In order that the desired incremental phase change can be obtained with only a moderate frequency scan, it is necessary to employ a large total phase shift (a long electrical path) in each feeder section. Since the free space distance between adjacent elements of the array is only of the order of one half wavelength it becomes necessary greatly to increase the electrical length of the feeder sections which excite adjacent elements in the array.
One way of doing this is to employ a tortuous feeder path, i.e., one with many bends (a serpentine). An alternative, and more sophisticated, method is to provide a feeder with a slow wave structure such as that occurring in a surface wave.
Existing serpentine structures are extremely complex and difficult to make, and existing surface wave structures are also difficult to make.
SUMMARY OF THE INVENTION
It is an object of the present invention to obtain a suitable slow wave structure in a simple and convenient way.
According to a broad aspect of the invention there is provided a frequency scanning waveguide antenna comprising a waveguide having a longitudinal axis, said waveguide being evanescent over the operating frequency band of said antenna; a plurality of capacitive elements within said waveguide each said element spaced apart from adjacent elements by a predetermined distance l and adjusted to provide a conjugate match condition for evanescent mode resonance; and a plurality of longitudinal radiating slots in the wall of said waveguide, each of said slots located midway between adjacent ones of said capacitive elements.
The invention will be better understood from the following description in conjunction with the accompanying drawings, in which:
BRIEF DESCRIPTION OF THE DRAWINGS
FIGS. 1 and 2 are schematic side and plan views respectively of a frequency scan waveguide antenna element with radiating slots;
FIG. 3 shows beam positions for the antenna of FIGS. 1 and 2;
FIG. 4 shows beam positions for an antenna having a different period of radiating slots;
FIG. 5 shows the measured beam angle as a function of frequency for the antenna of FIGS. 1 and 2;
FIG. 6 shows the measured polar diagram of the antenna for different frequencies; and
FIG. 7 shows a termination arrangement for the antenna.
DESCRIPTION OF THE PREFERRED EMBODIMENT
The waveguide antenna shown in FIGS. 1 and 2 comprises a multisection evanescent mode bandpass filter, basically as described in U.S. Pat. No. 3,621,483 constructed in rectangular cross-section waveguide 1 which is evanescent over the operating frequency band of the antenna. The waveguide contains capacitive elements, tuning screws 2, spaced apart by a distance l and adjusted to give the necessary conjugate match condition to achieve evanescent mode resonance for energy transfer along the waveguide over the required passband. Other forms of capacitive element may be used, for example, transverse slices of dielectric material.
Radiating longitudinal slots 3 are provided, as by milling, on alternate sides of the longitudinal center line of the waveguide either in the broad wall of the waveguide, as shown, or in the narrow wall. The transverse center line of each slot is midway between adjacent resonators (the screws 2).
When tuned, there is obtained a slow wave structure using a periodic network of evanescent mode resonators.
Image parameter theory is applicable to such networks analyzed on an impedance basis. The bandwidth of the network depends on the length, l, between resonators (the screws 2) and the propagation constant y.
The equations giving the image impedance, Z i , and the image transfer constant, θ, are given by
Z i = Z o √ tanh yl/2 [ Z o B 1 sinh yl/2 - cosh yl/2/sinh yl/2 - Z o B 1 cosh yl/2]
and
cosh θ = 2[ cosh 2 yl/2 - Z o B 1 sinh yl/2 cosh yl/2] - 1
The network will transmit energy freely over the frequency range in which its image impedance is real. The limits of this range occur at
cosh θ=±1.
Thus, provided -1 < cosh θ < 1, energy will be freely transmitted. For cosh θ to have values in this range, θ must be imaginary, i.e.,
θ=jβ
Thus, since cosh jβ=cos β=±1, at the band limit the limiting values for β are 0 and π, with a continuous range of variation in between.
The importance of the image transfer constant in this analysis lies in the fact that it gives the relationship between the voltages at the input and output of each section of the filter, (i.e., fields in the guide at a and b).
It will, therefore, be clear that as the frequency is varied over the transmission band of the filter the relative phase difference between planes a and b will vary smoothly between 0° and 180°, being approximately 90° at the band center. In a periodic structure such as the one shown, these phase differentials will be repeated at corresponding planes a 1 , b 1 ; a 2 , b 2 ; etc., throughout the structure.
To obtain a corresponding variation over a narrow band in a conventional straight propagating waveguide would necessitate a guide length 20-100 times the value required in evanescent waveguide functioning as described above.
The polar diagram of the antenna is determined by the relative phase and amplitude of the fields in the slots 3. If φ is the phase shift per slot period L then the beam angles Ψ p with respect to the normal of the array (X-axis) are given by the following equation for the structure shown in FIGS. 1 and 2.
sin Ψ p ± = τo/2L (φ/u ± 2p-1)
p = 1, 2, 3.
A real beam exists in direction Ψ p for each │ sin Ψ p │ ≤1. For L/τo = 1/√2 the beam positions are shown in FIG. 3.
Thus for this example at the lower cut-off frequency there exist two beams in space, Ψ 1 - at -45° and Ψ 1 + at 45°. As the frequency is increased to the upper cut-off frequency the beam Ψ 1 - moves to Ψ° while Ψ 1 + moves to +90° and then into imaginary space. There exists a region of about 25° (shown shaded) where Ψ 1 - is the only beam in space and for this slot period represents the useful range.
FIG. 4 shows the scan range over the passband for the example L/τo = 1/2.
At the lower cut-off frequency of the pass band there exists two beams in directions Ψ = ± 90°. As the frequency is increased Ψ 1 + moves beyond cut-off (into imaginary space), and Ψ 1 - increases and reaches Ψ° at the upper cut-off frequency. Thus in this example Ψ 1 - is the only beam in real space.
An antenna was made consisting of 19 periods with L/τo = 1/√2, and the measured beam angle of this antenna is shown as a function of frequency in FIG. 5. This linear characteristic has a slope equal to 27.6°/100 MHz at 5.7 GHz, i.e., 27.6° for 1.75 percent change of frequency.
The polar diagram is readily obtained once a knowledge of the field in the slots is known. For a structure having radiating slots each located on the same side of the longitudinal center line of the waveguide, with a period L, the beam angles are given by
sin Ψ = τo/2L (φ ± 2p)
p = 0, 1, 2, 3.
Typical measured polar diagrams are shown in FIG. 6, showing the change in beam angle with successive increases of frequency, each of 0.25 GHz, from 5.25 to 5.75 GHz, diagrams A, B and C respectively.
It will be noticed that in these polar diagrams there is an oppositely angled low-power beam. This arises because the antenna had resonators which were each of low loss capacitive elements (tuning screws), and the spurious beams resulted from energy being reflected from the far end of the antenna.
This unwanted effect is substantially eliminated by an appropriate load termination of the antenna. This is indicated in FIG. 7 by forming the final resonators with lossy capacitive elements, e.g., lossy tuning screws 2A. Typically the last three, or possibly five, resonators are so constituted, so that residual energy is progressively absorbed and not reflected. There are no radiating slots in this terminating portion of the antenna. The waveguide end is normally closed by a metal plate 4.
The antenna as above described possesses the following advantages:
a. uses commercially available waveguide,
b. the tuning elements introduced into the antenna compensate for the waveguide tolerances,
c. no complex machining process is necessary; only milling the slots,
d. is significantly lower in weight than the serpentine or surface wave antenna, and
e. uses a waveguide of width less than τo/2 which permits stacking of linear arrays for wide angle azimuth scanning.
It is to be understood that the foregoing description of specific examples of this invention is made by way of example only and is not to be considered as a limitation on its scope.