Title:
TEMPERATURE-STABLE DIELECTRIC RESONATOR FILTERS FOR STRIPLINE
United States Patent 3840828
Abstract:
There is disclosed a band-reject dielectric resonator filter for stripline, the resonators being disposed over the principal conductor and separated therefrom by a dielectric spacer and offset from the symmetry position over the stripline conductor. The offset is selected so that the magnetic field lines of the stripline pass through the planar, parallel surfaces of the resonator to a maximum degree. Large scale fabrication of the stripline is facilitated, while accurate normal positioning of the resonator is assured; and resonator to stripline coupling is relatively insensitive to small variations in the lateral position. Further, the principal conductor has no substantial degradation of the resonator Q. Strong coupling has been demonstrated with the use of low dielectric constant materials that are readily-temperature compensated. Also, disclosed are a plurality of such resonators coupled together to create specially-shaped reject bands or pass bands.
Inventors:
Linn, Donald Floyd (Kempton, PA)
Plourde, James Kevin (Allentown, PA)
Plourde, James Kevin (Allentown, PA)
Application Number:
05/413907
Publication Date:
10/08/1974
Filing Date:
11/08/1973
View Patent Images:
Export Citation:
Assignee:
Bell Telephone Laboratories, Incorporated (Murray Hill, NJ)
Primary Class:
Other Classes:
333/238, 333/204
International Classes:
H01P1/203; H01P7/10; H01P1/20; H01P7/00; H01P1/20; H01P3/08
Field of Search:
333/73S,83R,82B,82R,83T,84M,84R,97R,73R
Primary Examiner:
Lawrence, James W.
Assistant Examiner:
Nussbaum, Marvin
Attorney, Agent or Firm:
Wisner, Wilford L.
Claims:
What is claimed is
1. A microwave circuit comprising a stripline including a planar dielectric substrate and a principal conductor deposited on one surface of said substrate, and means for filtering a portion of the frequencies of microwave radiation transmitted through said stripline, comprising a dielectric resonator having planar surfaces parallel to the plane of said substrate and having a composition selected for temperature compensation of its resonant frequencies, and a dielectric spacer between said resonator and said conductor, said resonator being offset from symmetrical alignment over said conductor and adapted to support a TE01δ mode of its resonant frequencies, the degree of offset being selected to maximize the degree of coupling of said mode from said stripline to said resonator.
2. A microwave circuit according to claim 1 in which the composition of the dielectric resonator is selected to have positive coefficient of frequency variation with respect to temperature and the substrate of the stripline and housing effects are selected to have a negative coefficient of frequency variation with respect to temperature.
3. A microwave circuit according to claim 1 in which the center-to-center offset spacing of the dielectric resonator with respect to the principal conductor is substantially less than the half width of the dielectric resonator.
4. A microwave circuit according to claim 3 including a ground plane having a metallic housing structure about the dielectric resonator, the housing structure being selected to have the opposite coefficient frequency variation with respect to temperature as compared to that possessed by the dielectric resonator.
5. A microwave circuit according to claim 4 including means for tuning the filter characteristic of the dielectric resonator.
6. A microwave circuit according to claim 1 including a plurality of the dielectric resonators disposed at least partially over the principal conductor and separated by odd multiples of a quarter wavelengths from one another to minimize spurious coupling resonances therebetween, said circuit including a metallic housing structure intruding between said resonators.
7. A microwave circuit according to claim 1 in which the dielectric spacer is composed of a readily-machinable dielectric material, whereby the spacing between two of its surfaces may be accurately determined.
8. A microwave circuit according to claim 1 including a waveguide housing that is proportioned to be beyond cut-off for a band of frequencies to be propagated and including a plurality of dielectric resonators disposed within said housing with spacings for mutual coupling therebetween to provide propagation through the housing for the band of frequencies, the stripline comprising a planar dielectric substrate extending completely through the housing and a principal conductor that is broken in at least one region within said housing in proximity to one or more of the resonators.
9. A microwave circuit according to claim 8 in which the principal conductor is terminated under the end resonators of the plurality of resonators and is absent therebetween.
10. A microwave circuit according to claim 8 in which the plurality of dielectric resonators have spacings between adjacent resonators substantially equal to a quarterwave length for the center frequency of the band of frequencies and the principal conductor includes segments extending between adjacent resonators, so that the principal conductor lacks continuity only in a plurality of limited regions respectively adjacent to each of the resonators.
1. A microwave circuit comprising a stripline including a planar dielectric substrate and a principal conductor deposited on one surface of said substrate, and means for filtering a portion of the frequencies of microwave radiation transmitted through said stripline, comprising a dielectric resonator having planar surfaces parallel to the plane of said substrate and having a composition selected for temperature compensation of its resonant frequencies, and a dielectric spacer between said resonator and said conductor, said resonator being offset from symmetrical alignment over said conductor and adapted to support a TE01δ mode of its resonant frequencies, the degree of offset being selected to maximize the degree of coupling of said mode from said stripline to said resonator.
2. A microwave circuit according to claim 1 in which the composition of the dielectric resonator is selected to have positive coefficient of frequency variation with respect to temperature and the substrate of the stripline and housing effects are selected to have a negative coefficient of frequency variation with respect to temperature.
3. A microwave circuit according to claim 1 in which the center-to-center offset spacing of the dielectric resonator with respect to the principal conductor is substantially less than the half width of the dielectric resonator.
4. A microwave circuit according to claim 3 including a ground plane having a metallic housing structure about the dielectric resonator, the housing structure being selected to have the opposite coefficient frequency variation with respect to temperature as compared to that possessed by the dielectric resonator.
5. A microwave circuit according to claim 4 including means for tuning the filter characteristic of the dielectric resonator.
6. A microwave circuit according to claim 1 including a plurality of the dielectric resonators disposed at least partially over the principal conductor and separated by odd multiples of a quarter wavelengths from one another to minimize spurious coupling resonances therebetween, said circuit including a metallic housing structure intruding between said resonators.
7. A microwave circuit according to claim 1 in which the dielectric spacer is composed of a readily-machinable dielectric material, whereby the spacing between two of its surfaces may be accurately determined.
8. A microwave circuit according to claim 1 including a waveguide housing that is proportioned to be beyond cut-off for a band of frequencies to be propagated and including a plurality of dielectric resonators disposed within said housing with spacings for mutual coupling therebetween to provide propagation through the housing for the band of frequencies, the stripline comprising a planar dielectric substrate extending completely through the housing and a principal conductor that is broken in at least one region within said housing in proximity to one or more of the resonators.
9. A microwave circuit according to claim 8 in which the principal conductor is terminated under the end resonators of the plurality of resonators and is absent therebetween.
10. A microwave circuit according to claim 8 in which the plurality of dielectric resonators have spacings between adjacent resonators substantially equal to a quarterwave length for the center frequency of the band of frequencies and the principal conductor includes segments extending between adjacent resonators, so that the principal conductor lacks continuity only in a plurality of limited regions respectively adjacent to each of the resonators.
Description:
BACKGROUND OF THE INVENTION
This invention relates to filters for stripline in which dielectric resonators are used.
A variety of techniques has been employed for filtering in stripline circuits. Some of these, such as specific configurations of the copper of the principal conductor of the stripline, are limited in performance by the dissipation losses of the stripline resonators.
In the past several years a variety of discoveries relating to the use of dielectric resonator type filters for stripline have been made. It has been found generally desirable to use dielectric resonators in which the dielectric constant of the material, e.g., titanium dioxide (TiO 2 ), is at least about 100.
Unfortunately, such dielectric resonators experience an undue variation of their filtering properties with temperature change. In response to the need for a solution to the problem, relatively temperaturestable materials were discovered. These materials, such as ceramic barium titanite (Ba 2 Ti 9 O 20 ) or a composite resonator structure including lithium tantalate (LiTaO 2 ) have been used.
As pointed out in the July 1971 article by Tor Dag Iveland, "Dielectric Resonator Filters for Application in Microwave Integrated Circuits," IEEE Transactions on Microwave Theory and Techniques, Vol. MTT-19 page 643 to 644, the introduction of micro-integrated circuits has emphasized the need for high quality resonator elements for stripline with very small size so that the ground plane, or housing, configuration does not have to be unduly bulky for the integrated circuit geometry. Quoting the article, "the coupling mechanism is essentially based on the evanescent guide technique, but apart from Harrison's design, the coupling structure is kept in the plane of the substrate, containing both the filter and the connected circuits." The most recent description of Harrison's design is found in the article by A. Fox, "Temperature-Stable Low-Loss Microwave Filters Using Dielectric Resonators," Electronics Letters (GB), Vol. 8, page 582, Nov. 16, 1972. In that configuration the geometry is uniquely a cylindrical geometry. It is used primarily for coupling from coaxial cable to waveguide; and the only stripline portion thereof is in the coupling from the coaxial cable to the waveguide. The dielectric resonator is used as a part of the coupling arrangement.
It is therefore seen that, except for specialized applications, all dielectric resonator filters for stripline have been most conveniently located on the stripline substrate.
One problem associated with this configuration is that the degree of coupling is extremely sensitive to small variations in lateral positioning of the resonator. This fact makes it extremely difficult to place the resonator on the stripline as a part of the ordinary integrated circuit production technique. Moreover, integrated circuit techniques are impaired because of the substantial enlargement of the ground plane structure that is needed in the vicinity of the dielectric resonator.
SUMMARY OF THE INVENTION
According to our invention, the problems of accuracy of placement of a dielectric resonator with respect to a stripline are solved in a manner consistent with good temperature stability and overall compactness by positioning the dielectric resonator over the principal conductor of the stripline and providing precision spacing from the principal conductor in the normal direction by an accurately machined dielectric spacer.
It is one subsidiary feature of this novel configuration that an offset from the symmetry positioning over the stripline conductor is selected so that its magnetic field lines pass through the planar, parallel surfaces of the pillbox-shaped resonator to a maximum degree. Since the maximum coupling condition is characterized by a broad, flat maximum with respect to lateral positioning from the principal conductor, that coupling value is relatively insensitive to small variations in lateral position.
Advantageously, for relatively low-dielectric-constant resonators, the coupling is sufficiently strong and yields the coupling values required for typical microwave communications filters. Such materials are readily-temperature compensated. Such materials have recently become available with temperature-compensated characteristics. It is also found that the principal conductor has relatively small effect on the unloaded resonated Q since close coupling is achieved without a deleterisously close spacing between the dielectric resonator and the principal conductor. In other words, the dissipation losses due to the proximity of the principal conductor to the dielectric resonator are maintained at an acceptable level.
BRIEF DESCRIPTION OF THE DRAWING
Further features and advantages of our invention will become apparent from the following detailed description taken together with the drawing, in which:
FIG. 1 is a pictorial cross-sectional view of the typical prior art arrangement of dielectric resonators with respect to striplines;
FIG. 2 is a pictorial cross-sectional view of a stripline with ground plane and with a dielectric resonator arranged with respect thereto according to our invention;
FIG. 3 shows curves useful in explaining the characteristics of our invention;
FIG. 4 shows a modification of the embodiment of FIG. 2, using coupled multiple dielectric resonators shown in pictorial cross-sectional view along the stripline;
FIG. 5 shows a plan view of the embodiment of FIG. 4;
FIGS. 6, 7 and 8 show the filter characteristic for the embodiment of FIGS. 4 and 5;
FIGS. 9A and 9B show pictorial plan and elevation views of a pass-band filter employing the invention;
FIGS. 10A and 10B show a modification of the embodiment of FIGS. 9A and 9B;
FIGS. 11 and 13 show curves useful in explaining the embodiment of FIGS. 9A and 9B; and
FIG. 12 is a partial showing of FIG. 9A and setting forth parameters used in the curves of FIG. 11.
DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS
In the prior art embodiment shown in FIG. 1 which has been previously characterized as the typical prior art, the dielectric resonator 11 is coupled to the stripline 13 by positioning it on the substrate 12 next to the principal conductor 14. The stripline 13 also includes the dielectric layers 15 and 16, which may be air where desired, and the ground plane 17. With the use of relatively low dielectric constant resonators, e.g., ε.about. 50, the stripline housing width becomes excessive and allows spurious waveguide type modes to propagate, resulting in undesirable spurious filter responses.
A further drawback found with the arrangement of FIG. 1 is that the dielectric resonator 11 is not located at the maximum-coupling position with respect to the principal conductor; and the degree of coupling is extraordinarily sensitive to very minute variations in the lateral center-to-center spacing d between the principal conductors 14 and the resonator 11. In contrast, an arrangement of a dielectric resonator 21 with respect to a stripline 23 as shown in FIG. 2 overcomes the foregoing problems by disposing the dielectric resonator 21 over the principal conductor 24 of the stripline so that in addition to a lateral center-to-center space d the resonator positioning is now characterized by a normal center-to-center spacing h. This normal positioning is determined by a dielectric spacer 25, which may be accurately machined in a separate operation from that of positioning the resonator 21 over the substrate 22.
For completeness of illustration the stripline 23 is shown with its substrate 22 and the additional layer of dielectric material 26 thereunder for structural rigidity. Dielectric layer 26 may be omitted where desired. Typically, the ground plane 27 in the vicinity of the resonator 21 encloses the resonator within a metal housing, similar to a section of waveguide, through which a tuning screw 28 is readily inserted.
It is particularly important in the embodiment of our invention shown in FIG. 2 that the spacing parameter d is selected for maximum coupling. In particular, the coupling is absolutely a maximum with respect to the lateral offset d and is relatively insensitive to small variations in d because the maximum is a broad flat maximum. Illustratively, the increased top ground plane clearance shown in FIG. 2 is provided in the vicinity of dielectric resonators only so that spurious housing resonances are eliminated. In other words, the same characteristic impedance as in the preceding and following stripline portions should be maintained throughout the resonator region, for frequencies outside the reject band of the filter.
Illustratively, the width of the principal conductor 24 will be altered in the vicinity of resonators such as resonator 21 in order to provide an impedance match along the line at these points and thus avoid spurious reflections along the stripline.
We have found that relatively low dielectric constant resonators (ε≉38) can be used in this configuration. One exemplary embodiment employed barium titanate (Ba 2 Ti9O 20 ).
It is expected that the configuration of FIG. 2 will be found very useful in radio relay applications and similar applications, where its reduced size and weight as compared to alternative filters will be advantageous.
The band reject filter comprising dielectric resonator 21 utilizes either a composition material of lithium tantalate and titanium dioxide or, alternatively, uses barium titanate (Ba 2 Ti 9 O 20 ). The use of a lithium tantalate and titanium dioxide composite is disclosed in some detail in the copending patent application of one of us, J. K. Plourde, Ser. No. 317,385 filed Dec. 22, 1972, assigned to the assignee hereof. The use of Ba 2 Ti 9 O 20 is disclosed in the copending patent application of H. M. O'Bryan-J. K. Plourde-J. Thomson, Jr. Ser. No. 394,187 filed Sept. 4, 1973 and assigned to the assignee hereof.
In the operation of the embodiments of FIG. 2 between 3.7 and 4.2 GHZ an out-of-band return loss greater than 25 dB was obtained; and spurious or waveguide type mode resonances in the stripline housing or ground plane 27 were eliminated.
Although higher order modes or multiple resonator modes may possibly be used to advantage for particular filter applications, only the lowest order cylindrical mode, the TE 01 δ, was considered in our early embodiments.
In FIG. 3, there are shown curves 31, 32 and 33 of the resonator external Q with respect to the lateral offset spacing d for various values of the normal spacing h. The desirable value of d is in every case at the low part of the curve which corresponds to maximum coupling between the principal conductor 24 and the resonator 21. It will be noted that the resonator and therefore the coupling is insensitive to small variations in d about the preferred value. Because of this fact, the tolerance in d can be relaxed; and only the value h need be accurately controlled. The precise control in the value of h is easily obtained by positioning the resonator 21 on a dielectric spacer 25, such as a Rexolite disc. The height of the resonator 21 was illustratively 0.175 inch and its radius in the horizontal plane was 0.312 inch. It will be noted that, whereas in the prior art configuration the spacing d is significantly greater than the resonator radius, in our configuration the spacing d can be significantly less than the resonator radius with a corresponding reduction of the required lateral dimensions of the housing 27 and a reduction in spurious resonances.
For a multiple-resonator bandstop filter as shown in FIG. 4, the spacing between dielectric resonators 41, 51 and 61 is an odd multiple of a quarter wavelength. The minimum allowable resonator spacing will be the spacing for which the spurious coupling between resonators is held to an acceptable level and is determined by the excitation level of the higher order modes. Advantageously, a housing possessing small, cross-sectional dimensions, according to our invention, will permit the resonators to be closely spaced. Illustratively, each resonator 41, 51 and 61 is provided with its own tuning screw 48, 58, and 68. The entire assembly 43 is shown with the coaxial couplings 69 and 70. In FIG. 5 a plane view in section showing principally the configuration of the principal conductor 44 illustrates where the impedance-matching changes in width of the principal conductor occur with respect to the lateral dimensions of the resonators 41, 51 and 61. Specifically, these changes in width are vertically aligned with the resonators. The tuning screws 48, 58 and 68 allow plus (+) frequency tuning when the screw is a conductor or minus (-) frequency tuning when the screw is a dielectric. In other respects the design and operation of the filters of FIG. 4 and 5 follow known principles for dielectric resonators used with stripline.
Some typical filter characteristics are shown in FIGS. 6-8. The curves 71, 81 generally show the return loss │S 11 │ versus frequency; the curves 72, 82 show transmission │S 21 │ versus frequency.
Characteristics for a filter utilizing Ba 2 Ti 9 O 20 resonators are represented in FIG. 6. The unloaded resonator Q, Q o , can be calculated giving an approximate value of 6,300.
The above Q o value compares well with the undegraded resonator Q of 6,780 verifying that the Q degradation is not significant with this filter configuration. The out-of-band return loss over 3.7 to 4.2 GHz is 26dB. This parameter is determined by the uniformity of the stripline impedance throughout the filter rather than upon the resonator properties themselves. The peak insertion loss is approximately given by the following equation: ##SPC1##
Equation 1 gives 104 dB whereas the measured value is 84 dB.
Curves 81 and 82, │S 11 │ and │S 21 │ versus frequency for a filter utilizing LiTaO 3 /TiO 2 composite resonators are given in FIG. 7. Q o = 2,800, showing some Q o degradation from the undegraded resonator Q of 3,820. The coupling Q, Q c , (1/Q c = (1/Q omeas ) - 1/Q o ), i.e., the Q associated with the dissipation losses external to the resonators is 10,463. The out-of-band return loss is measured at 28 dB. Equation 2 yields a peak insertion loss of 91 dB whereas the measured value is 64 dB. The out-of-band insertion loss for the Ba 2 Ti 9 O 20 and LiTaO 3 /TiO 2 filters is 0.10 dB and 0.175 dB, respectively. This parameter is essentially independent of the dielectric resonator properties and depends upon the quality of the transmission line used in the filter.
In FIG. 8 the curve 91 shows the transmission characteristic of the multiple resonator filter versus frequency at 40° Fahrenheit. The same transmission versus frequency characteristic is shown by the displaced curve 92 for a temperature of 141° Fahrenheit. The shift then characteristically was 2.625 MHz. This value is less than the 3.17 MHz value shift predicted from the temperature coefficient of an isolated resonator because of a small compensating effect related to the use of an alumina substrate 42 and the metal filter housing 47, for the case of use of barium titanate in the resonators 41, 51 and 61. The temperature coefficient of frequency is positive for barium titanate and is negative for alumina and the housing effect. The resonators used in this filter possessed an average temperature coefficient of frequency, τ f = + 14.3 ppm°C. Similar materials for dielectric resonators can be made to yield temperature coefficients as low as τ f = 2.5 ppm°C. A practical tolerance range would be τ f = 0 ± 5 ppm/°C with dielectric resonator tolerances dominating. Such filters are essentially temperature compensated. It is found that the abovedescribed filter characteristics are far superior to those of a typical copper comb-type filter used with stripline.
While it is well known for other types of transmission media and other resonator-filters that a band-pass filter can be constructed by modifying a bandstop filter, it is instructive to consider how those principles are applicable to the present invention. A band-pass filter is formed by positioning one or more resonators in a section of waveguide that is beyond cut-off at the frequencies of interest, in the absence of the resonators.
In the illustrative band-pass filter of FIG. 9A, the dielectric resonators 103, which are like resonator 21 of FIG. 2 or respectively like resonators 41, 51 and 61 of FIGS. 4 and 5, are spaced apart in housing 101 above stripline 102.
The housing 101 is a section of waveguide, illustratively rectangular, which is beyond cut-off; that is, it will not propagate the frequencies of the intended pass-band in the absence of resonators 103. The intended pass-band frequencies lie in a band which is centered at the resonant frequencies of the resonators 103. Electromagnetic energy is coupled through the structure from left to right in the drawing. At other frequencies, outside of the pass-band, very little energy is propagated through the structure.
While both direct-coupled and quarterwave-coupled band-pass filters will be described, FIGS. 9A and 9B show the direct-coupled band-pass filter. The input and output sections of stripline 102 are coupled to the end resonators 103 and the inner ones of the resonators 103 directly couple energy to adjacent resonators. The resonators couplings and hence the filter characteristics are determined by the inter-resonator spacings between the inner resonators as well as the coupling between the end resonators and the input and output striplines.
As in the preceding embodiments of the invention, the resonators 103 are disposed over the stripline 102 in the position that is basically off center with respect to at least portions 109 and 110 of the principal conductor 105 of the stripline 102. The reduction in principal conductor width illustrated by portions 109 and 110 serve the dual purpose of providing appropriate impedance matching to the resonators, as described above, as well as facilitating the off-center spacing. The resonators 103 are spaced from the principal conductor 105 of the stripline 102 and from portions 109 and 110 by dielectric spacers, the dielectric spacer 106 being over portion 109, the spacer 107 being over a portion of stripline 102 having no principal conductor within the interior of the housing 101, and the spacer 108 being over the portion 110 of stripline 102. The lateral distance between the resonator and the end of the stripline principle conductor is determined such that strong coupling is obtained between the stripline and the end resonator while providing negligible coupling between the stripline and the inner resonators. These spacers may be made as explained for the preceding embodiments.
The direct-coupled band-pass filter of FIGS. 9A and 9B typically yields the best results of the types we have investigated and, in addition, is smaller and simpler than the quarterwave-coupled type described hereinafter.
The coupling characteristics between an end resonator and the stripline are set forth in FIG. 11 by curves 121 through 124, in terms of the effective overall, Q ex , of the filter as a function of the spacing A from the resonator edge to principal conductor edge, as shown in FIG. 12. The respective curves 121 through 124 represent the differing spacer thicknesses of spacers 106 through 108 for different filter designs, specifically 0.160 inch, 0.130 inch, 0.120 inch and 0.060 inch for all three like spacers for the respective different designs.
Except for the fact that the stripline 102 is terminated, that is open circuited, near or beyond the end resonators 103, the coupling is similar to that used in our band-elimination filters described above in that the lateral offset is adjusted to the position yielding a broad, flat maximum of coupling to the resonant mode of the resonator to be utilized. Most of the fabrication advantages of our invention and other advantages described above also apply here for that reason.
For the sake of completeness we list the features of the band-pass filters of FIGS. 9A and 9B as follows:
1. Strong coupling, with low Q ex , is obtainable between the dielectric resonator and the stripline.
2. The coupling can be precisely controlled and is a function of the thickness of the dielectric spacers, for example spacers 106 through 108, used to locate the resonators 103 over the stripline.
3. The coupling is relatively insensitive to the lateral off-set with respect to the center conductor.
4. The spacing of the resonators over the center conductor reduces the degradation of the resonator Q due to conductor loss in the bottom ground plane of housing 101.
5. The clearance from the resonator to the top ground plane is sufficient to limit the resonator Q degradation to an acceptable level.
6. The housing width is reduced such that spurious housing resonances are eliminated.
7. Relatively low dielectric constant, ε≉39, materials can be used.
A quarterwave-coupled band-pass filter is illustrated in FIGS. 10A and 10B. This structure resembles a number of single resonator filters connected in cascade and spaced from one another by odd multiples of a quarterwave length. The filter of FIGS. 10A and 10B can be compared to that of FIGS. 9A and 9B in that the spacing between resonators 103 is about three-fourths of a wavelength between each pair and in that sections of the principal conductor 105 extend between the resonators so that the only portion of stripline 102 free of principal conductor is a small space directly under each of the resonators 103.
This configuration is larger and more difficult to fabricate than that of FIGS. 9A and 9B.
The filter characteristics may be determined by the coupling to each resonator.
The performance of a direct-coupled band-pass filter utilizing three resonators is presented in FIG. 13.
It should be noted that, on the vertical axis in FIG. 13, │S 21 │ represents the absolute value of transmission and │S 11 │ represents the absolute value of return loss. The 35 dB return loss of curve 132 at fo compares very favorably with that of other band-pass filters. The 0.4 dB insertion loss of curve 131 at fo corresponds to a resonator Q of 3,260. The undegraded resonator Q is approximately 6,000.
A major contribution to this Q degradation is made by both the adhesive used to bond the resonators in the circuit and the metallic tuning screws (not shown) and can be reduced by further optimization where desired. No spurious transmission responses are observed in the 3.7 to 4.2 GHz band. The first spurious response occurs at approximately 4.6GHz and is caused by a higher order resonator mode.
This invention relates to filters for stripline in which dielectric resonators are used.
A variety of techniques has been employed for filtering in stripline circuits. Some of these, such as specific configurations of the copper of the principal conductor of the stripline, are limited in performance by the dissipation losses of the stripline resonators.
In the past several years a variety of discoveries relating to the use of dielectric resonator type filters for stripline have been made. It has been found generally desirable to use dielectric resonators in which the dielectric constant of the material, e.g., titanium dioxide (TiO 2 ), is at least about 100.
Unfortunately, such dielectric resonators experience an undue variation of their filtering properties with temperature change. In response to the need for a solution to the problem, relatively temperaturestable materials were discovered. These materials, such as ceramic barium titanite (Ba 2 Ti 9 O 20 ) or a composite resonator structure including lithium tantalate (LiTaO 2 ) have been used.
As pointed out in the July 1971 article by Tor Dag Iveland, "Dielectric Resonator Filters for Application in Microwave Integrated Circuits," IEEE Transactions on Microwave Theory and Techniques, Vol. MTT-19 page 643 to 644, the introduction of micro-integrated circuits has emphasized the need for high quality resonator elements for stripline with very small size so that the ground plane, or housing, configuration does not have to be unduly bulky for the integrated circuit geometry. Quoting the article, "the coupling mechanism is essentially based on the evanescent guide technique, but apart from Harrison's design, the coupling structure is kept in the plane of the substrate, containing both the filter and the connected circuits." The most recent description of Harrison's design is found in the article by A. Fox, "Temperature-Stable Low-Loss Microwave Filters Using Dielectric Resonators," Electronics Letters (GB), Vol. 8, page 582, Nov. 16, 1972. In that configuration the geometry is uniquely a cylindrical geometry. It is used primarily for coupling from coaxial cable to waveguide; and the only stripline portion thereof is in the coupling from the coaxial cable to the waveguide. The dielectric resonator is used as a part of the coupling arrangement.
It is therefore seen that, except for specialized applications, all dielectric resonator filters for stripline have been most conveniently located on the stripline substrate.
One problem associated with this configuration is that the degree of coupling is extremely sensitive to small variations in lateral positioning of the resonator. This fact makes it extremely difficult to place the resonator on the stripline as a part of the ordinary integrated circuit production technique. Moreover, integrated circuit techniques are impaired because of the substantial enlargement of the ground plane structure that is needed in the vicinity of the dielectric resonator.
SUMMARY OF THE INVENTION
According to our invention, the problems of accuracy of placement of a dielectric resonator with respect to a stripline are solved in a manner consistent with good temperature stability and overall compactness by positioning the dielectric resonator over the principal conductor of the stripline and providing precision spacing from the principal conductor in the normal direction by an accurately machined dielectric spacer.
It is one subsidiary feature of this novel configuration that an offset from the symmetry positioning over the stripline conductor is selected so that its magnetic field lines pass through the planar, parallel surfaces of the pillbox-shaped resonator to a maximum degree. Since the maximum coupling condition is characterized by a broad, flat maximum with respect to lateral positioning from the principal conductor, that coupling value is relatively insensitive to small variations in lateral position.
Advantageously, for relatively low-dielectric-constant resonators, the coupling is sufficiently strong and yields the coupling values required for typical microwave communications filters. Such materials are readily-temperature compensated. Such materials have recently become available with temperature-compensated characteristics. It is also found that the principal conductor has relatively small effect on the unloaded resonated Q since close coupling is achieved without a deleterisously close spacing between the dielectric resonator and the principal conductor. In other words, the dissipation losses due to the proximity of the principal conductor to the dielectric resonator are maintained at an acceptable level.
BRIEF DESCRIPTION OF THE DRAWING
Further features and advantages of our invention will become apparent from the following detailed description taken together with the drawing, in which:
FIG. 1 is a pictorial cross-sectional view of the typical prior art arrangement of dielectric resonators with respect to striplines;
FIG. 2 is a pictorial cross-sectional view of a stripline with ground plane and with a dielectric resonator arranged with respect thereto according to our invention;
FIG. 3 shows curves useful in explaining the characteristics of our invention;
FIG. 4 shows a modification of the embodiment of FIG. 2, using coupled multiple dielectric resonators shown in pictorial cross-sectional view along the stripline;
FIG. 5 shows a plan view of the embodiment of FIG. 4;
FIGS. 6, 7 and 8 show the filter characteristic for the embodiment of FIGS. 4 and 5;
FIGS. 9A and 9B show pictorial plan and elevation views of a pass-band filter employing the invention;
FIGS. 10A and 10B show a modification of the embodiment of FIGS. 9A and 9B;
FIGS. 11 and 13 show curves useful in explaining the embodiment of FIGS. 9A and 9B; and
FIG. 12 is a partial showing of FIG. 9A and setting forth parameters used in the curves of FIG. 11.
DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS
In the prior art embodiment shown in FIG. 1 which has been previously characterized as the typical prior art, the dielectric resonator 11 is coupled to the stripline 13 by positioning it on the substrate 12 next to the principal conductor 14. The stripline 13 also includes the dielectric layers 15 and 16, which may be air where desired, and the ground plane 17. With the use of relatively low dielectric constant resonators, e.g., ε.about. 50, the stripline housing width becomes excessive and allows spurious waveguide type modes to propagate, resulting in undesirable spurious filter responses.
A further drawback found with the arrangement of FIG. 1 is that the dielectric resonator 11 is not located at the maximum-coupling position with respect to the principal conductor; and the degree of coupling is extraordinarily sensitive to very minute variations in the lateral center-to-center spacing d between the principal conductors 14 and the resonator 11. In contrast, an arrangement of a dielectric resonator 21 with respect to a stripline 23 as shown in FIG. 2 overcomes the foregoing problems by disposing the dielectric resonator 21 over the principal conductor 24 of the stripline so that in addition to a lateral center-to-center space d the resonator positioning is now characterized by a normal center-to-center spacing h. This normal positioning is determined by a dielectric spacer 25, which may be accurately machined in a separate operation from that of positioning the resonator 21 over the substrate 22.
For completeness of illustration the stripline 23 is shown with its substrate 22 and the additional layer of dielectric material 26 thereunder for structural rigidity. Dielectric layer 26 may be omitted where desired. Typically, the ground plane 27 in the vicinity of the resonator 21 encloses the resonator within a metal housing, similar to a section of waveguide, through which a tuning screw 28 is readily inserted.
It is particularly important in the embodiment of our invention shown in FIG. 2 that the spacing parameter d is selected for maximum coupling. In particular, the coupling is absolutely a maximum with respect to the lateral offset d and is relatively insensitive to small variations in d because the maximum is a broad flat maximum. Illustratively, the increased top ground plane clearance shown in FIG. 2 is provided in the vicinity of dielectric resonators only so that spurious housing resonances are eliminated. In other words, the same characteristic impedance as in the preceding and following stripline portions should be maintained throughout the resonator region, for frequencies outside the reject band of the filter.
Illustratively, the width of the principal conductor 24 will be altered in the vicinity of resonators such as resonator 21 in order to provide an impedance match along the line at these points and thus avoid spurious reflections along the stripline.
We have found that relatively low dielectric constant resonators (ε≉38) can be used in this configuration. One exemplary embodiment employed barium titanate (Ba 2 Ti9O 20 ).
It is expected that the configuration of FIG. 2 will be found very useful in radio relay applications and similar applications, where its reduced size and weight as compared to alternative filters will be advantageous.
The band reject filter comprising dielectric resonator 21 utilizes either a composition material of lithium tantalate and titanium dioxide or, alternatively, uses barium titanate (Ba 2 Ti 9 O 20 ). The use of a lithium tantalate and titanium dioxide composite is disclosed in some detail in the copending patent application of one of us, J. K. Plourde, Ser. No. 317,385 filed Dec. 22, 1972, assigned to the assignee hereof. The use of Ba 2 Ti 9 O 20 is disclosed in the copending patent application of H. M. O'Bryan-J. K. Plourde-J. Thomson, Jr. Ser. No. 394,187 filed Sept. 4, 1973 and assigned to the assignee hereof.
In the operation of the embodiments of FIG. 2 between 3.7 and 4.2 GHZ an out-of-band return loss greater than 25 dB was obtained; and spurious or waveguide type mode resonances in the stripline housing or ground plane 27 were eliminated.
Although higher order modes or multiple resonator modes may possibly be used to advantage for particular filter applications, only the lowest order cylindrical mode, the TE 01 δ, was considered in our early embodiments.
In FIG. 3, there are shown curves 31, 32 and 33 of the resonator external Q with respect to the lateral offset spacing d for various values of the normal spacing h. The desirable value of d is in every case at the low part of the curve which corresponds to maximum coupling between the principal conductor 24 and the resonator 21. It will be noted that the resonator and therefore the coupling is insensitive to small variations in d about the preferred value. Because of this fact, the tolerance in d can be relaxed; and only the value h need be accurately controlled. The precise control in the value of h is easily obtained by positioning the resonator 21 on a dielectric spacer 25, such as a Rexolite disc. The height of the resonator 21 was illustratively 0.175 inch and its radius in the horizontal plane was 0.312 inch. It will be noted that, whereas in the prior art configuration the spacing d is significantly greater than the resonator radius, in our configuration the spacing d can be significantly less than the resonator radius with a corresponding reduction of the required lateral dimensions of the housing 27 and a reduction in spurious resonances.
For a multiple-resonator bandstop filter as shown in FIG. 4, the spacing between dielectric resonators 41, 51 and 61 is an odd multiple of a quarter wavelength. The minimum allowable resonator spacing will be the spacing for which the spurious coupling between resonators is held to an acceptable level and is determined by the excitation level of the higher order modes. Advantageously, a housing possessing small, cross-sectional dimensions, according to our invention, will permit the resonators to be closely spaced. Illustratively, each resonator 41, 51 and 61 is provided with its own tuning screw 48, 58, and 68. The entire assembly 43 is shown with the coaxial couplings 69 and 70. In FIG. 5 a plane view in section showing principally the configuration of the principal conductor 44 illustrates where the impedance-matching changes in width of the principal conductor occur with respect to the lateral dimensions of the resonators 41, 51 and 61. Specifically, these changes in width are vertically aligned with the resonators. The tuning screws 48, 58 and 68 allow plus (+) frequency tuning when the screw is a conductor or minus (-) frequency tuning when the screw is a dielectric. In other respects the design and operation of the filters of FIG. 4 and 5 follow known principles for dielectric resonators used with stripline.
Some typical filter characteristics are shown in FIGS. 6-8. The curves 71, 81 generally show the return loss │S 11 │ versus frequency; the curves 72, 82 show transmission │S 21 │ versus frequency.
Characteristics for a filter utilizing Ba 2 Ti 9 O 20 resonators are represented in FIG. 6. The unloaded resonator Q, Q o , can be calculated giving an approximate value of 6,300.
The above Q o value compares well with the undegraded resonator Q of 6,780 verifying that the Q degradation is not significant with this filter configuration. The out-of-band return loss over 3.7 to 4.2 GHz is 26dB. This parameter is determined by the uniformity of the stripline impedance throughout the filter rather than upon the resonator properties themselves. The peak insertion loss is approximately given by the following equation: ##SPC1##
Equation 1 gives 104 dB whereas the measured value is 84 dB.
Curves 81 and 82, │S 11 │ and │S 21 │ versus frequency for a filter utilizing LiTaO 3 /TiO 2 composite resonators are given in FIG. 7. Q o = 2,800, showing some Q o degradation from the undegraded resonator Q of 3,820. The coupling Q, Q c , (1/Q c = (1/Q omeas ) - 1/Q o ), i.e., the Q associated with the dissipation losses external to the resonators is 10,463. The out-of-band return loss is measured at 28 dB. Equation 2 yields a peak insertion loss of 91 dB whereas the measured value is 64 dB. The out-of-band insertion loss for the Ba 2 Ti 9 O 20 and LiTaO 3 /TiO 2 filters is 0.10 dB and 0.175 dB, respectively. This parameter is essentially independent of the dielectric resonator properties and depends upon the quality of the transmission line used in the filter.
In FIG. 8 the curve 91 shows the transmission characteristic of the multiple resonator filter versus frequency at 40° Fahrenheit. The same transmission versus frequency characteristic is shown by the displaced curve 92 for a temperature of 141° Fahrenheit. The shift then characteristically was 2.625 MHz. This value is less than the 3.17 MHz value shift predicted from the temperature coefficient of an isolated resonator because of a small compensating effect related to the use of an alumina substrate 42 and the metal filter housing 47, for the case of use of barium titanate in the resonators 41, 51 and 61. The temperature coefficient of frequency is positive for barium titanate and is negative for alumina and the housing effect. The resonators used in this filter possessed an average temperature coefficient of frequency, τ f = + 14.3 ppm°C. Similar materials for dielectric resonators can be made to yield temperature coefficients as low as τ f = 2.5 ppm°C. A practical tolerance range would be τ f = 0 ± 5 ppm/°C with dielectric resonator tolerances dominating. Such filters are essentially temperature compensated. It is found that the abovedescribed filter characteristics are far superior to those of a typical copper comb-type filter used with stripline.
While it is well known for other types of transmission media and other resonator-filters that a band-pass filter can be constructed by modifying a bandstop filter, it is instructive to consider how those principles are applicable to the present invention. A band-pass filter is formed by positioning one or more resonators in a section of waveguide that is beyond cut-off at the frequencies of interest, in the absence of the resonators.
In the illustrative band-pass filter of FIG. 9A, the dielectric resonators 103, which are like resonator 21 of FIG. 2 or respectively like resonators 41, 51 and 61 of FIGS. 4 and 5, are spaced apart in housing 101 above stripline 102.
The housing 101 is a section of waveguide, illustratively rectangular, which is beyond cut-off; that is, it will not propagate the frequencies of the intended pass-band in the absence of resonators 103. The intended pass-band frequencies lie in a band which is centered at the resonant frequencies of the resonators 103. Electromagnetic energy is coupled through the structure from left to right in the drawing. At other frequencies, outside of the pass-band, very little energy is propagated through the structure.
While both direct-coupled and quarterwave-coupled band-pass filters will be described, FIGS. 9A and 9B show the direct-coupled band-pass filter. The input and output sections of stripline 102 are coupled to the end resonators 103 and the inner ones of the resonators 103 directly couple energy to adjacent resonators. The resonators couplings and hence the filter characteristics are determined by the inter-resonator spacings between the inner resonators as well as the coupling between the end resonators and the input and output striplines.
As in the preceding embodiments of the invention, the resonators 103 are disposed over the stripline 102 in the position that is basically off center with respect to at least portions 109 and 110 of the principal conductor 105 of the stripline 102. The reduction in principal conductor width illustrated by portions 109 and 110 serve the dual purpose of providing appropriate impedance matching to the resonators, as described above, as well as facilitating the off-center spacing. The resonators 103 are spaced from the principal conductor 105 of the stripline 102 and from portions 109 and 110 by dielectric spacers, the dielectric spacer 106 being over portion 109, the spacer 107 being over a portion of stripline 102 having no principal conductor within the interior of the housing 101, and the spacer 108 being over the portion 110 of stripline 102. The lateral distance between the resonator and the end of the stripline principle conductor is determined such that strong coupling is obtained between the stripline and the end resonator while providing negligible coupling between the stripline and the inner resonators. These spacers may be made as explained for the preceding embodiments.
The direct-coupled band-pass filter of FIGS. 9A and 9B typically yields the best results of the types we have investigated and, in addition, is smaller and simpler than the quarterwave-coupled type described hereinafter.
The coupling characteristics between an end resonator and the stripline are set forth in FIG. 11 by curves 121 through 124, in terms of the effective overall, Q ex , of the filter as a function of the spacing A from the resonator edge to principal conductor edge, as shown in FIG. 12. The respective curves 121 through 124 represent the differing spacer thicknesses of spacers 106 through 108 for different filter designs, specifically 0.160 inch, 0.130 inch, 0.120 inch and 0.060 inch for all three like spacers for the respective different designs.
Except for the fact that the stripline 102 is terminated, that is open circuited, near or beyond the end resonators 103, the coupling is similar to that used in our band-elimination filters described above in that the lateral offset is adjusted to the position yielding a broad, flat maximum of coupling to the resonant mode of the resonator to be utilized. Most of the fabrication advantages of our invention and other advantages described above also apply here for that reason.
For the sake of completeness we list the features of the band-pass filters of FIGS. 9A and 9B as follows:
1. Strong coupling, with low Q ex , is obtainable between the dielectric resonator and the stripline.
2. The coupling can be precisely controlled and is a function of the thickness of the dielectric spacers, for example spacers 106 through 108, used to locate the resonators 103 over the stripline.
3. The coupling is relatively insensitive to the lateral off-set with respect to the center conductor.
4. The spacing of the resonators over the center conductor reduces the degradation of the resonator Q due to conductor loss in the bottom ground plane of housing 101.
5. The clearance from the resonator to the top ground plane is sufficient to limit the resonator Q degradation to an acceptable level.
6. The housing width is reduced such that spurious housing resonances are eliminated.
7. Relatively low dielectric constant, ε≉39, materials can be used.
A quarterwave-coupled band-pass filter is illustrated in FIGS. 10A and 10B. This structure resembles a number of single resonator filters connected in cascade and spaced from one another by odd multiples of a quarterwave length. The filter of FIGS. 10A and 10B can be compared to that of FIGS. 9A and 9B in that the spacing between resonators 103 is about three-fourths of a wavelength between each pair and in that sections of the principal conductor 105 extend between the resonators so that the only portion of stripline 102 free of principal conductor is a small space directly under each of the resonators 103.
This configuration is larger and more difficult to fabricate than that of FIGS. 9A and 9B.
The filter characteristics may be determined by the coupling to each resonator.
The performance of a direct-coupled band-pass filter utilizing three resonators is presented in FIG. 13.
It should be noted that, on the vertical axis in FIG. 13, │S 21 │ represents the absolute value of transmission and │S 11 │ represents the absolute value of return loss. The 35 dB return loss of curve 132 at fo compares very favorably with that of other band-pass filters. The 0.4 dB insertion loss of curve 131 at fo corresponds to a resonator Q of 3,260. The undegraded resonator Q is approximately 6,000.
A major contribution to this Q degradation is made by both the adhesive used to bond the resonators in the circuit and the metallic tuning screws (not shown) and can be reduced by further optimization where desired. No spurious transmission responses are observed in the 3.7 to 4.2 GHz band. The first spurious response occurs at approximately 4.6GHz and is caused by a higher order resonator mode.