Title:
FILTER-TYPE DIGITAL DIODE PHASE SHIFTER
United States Patent 3778733
Abstract:
Three PIN diodes form a T junction. When the series diodes are forward biased, the shunt diode is reverse-biased to effectively form a low pass circuit. When the series diodes are reverse-biased, the shunt diodes are forward-biased to form effectively a high pass circuit. The cut off frequency of the low and high pass states are above and below, respectively, the operating band so that each T section selectively introduces a phase delay and advance in the low and high pass states, respectively.
US Patent References:
MICROWAVE PHASE SHIFTER
DiPiazza - December 1970 - 3546636
High frequency solid state switch employing diodes with shiftable bias to control signal transmission
Spallone - August 1964 - 3146357
Controllable phase shift circuit
Woster - December 1967 - 3356865
CONTINUOUSLY VARIABLE HIGH-FREQUENCY TRANSMISSION LINE ATTENUATOR USING VARIABLY BIASED MICROWAVE DIODES AND METHOD THEREFOR
Russell - July 1969 - 3453564
MICROWAVE PHASE SHIFTER
DiPiazza - December 1970 - 3546636
High frequency solid state switch employing diodes with shiftable bias to control signal transmission
Spallone - August 1964 - 3146357
Controllable phase shift circuit
Woster - December 1967 - 3356865
CONTINUOUSLY VARIABLE HIGH-FREQUENCY TRANSMISSION LINE ATTENUATOR USING VARIABLY BIASED MICROWAVE DIODES AND METHOD THEREFOR
Russell - July 1969 - 3453564
Application Number:
05/251116
Publication Date:
12/11/1973
Filing Date:
05/08/1972
View Patent Images:
Export Citation:
Assignee:
Alpha Industries, Inc. (Woburn, MA)
Primary Class:
International Classes:
H01P1/185; H01P1/18; H01P1/18
Field of Search:
333/31R,7R,73R,73C,73S 307/256,259
Primary Examiner:
Gensler, Paul L.
Claims:
What is claimed is
1. Microwave phase shifting apparatus comprising,
1. Microwave phase shifting apparatus comprising,
Description:
BACKGROUND OF THE INVENTION
The present invention relates in general to phase shifting and more particularly concerns a novel microwave phase shifter that advantageously employs PIN diodes to provide a reliable digital phase shifter with relatively compact inexpensive reliable apparatus. The invention realizes large phase changes at microwave frequencies in a very compact length.
It is an important object of the invention to provide an improved phase shifter.
It is another object of the invention to achieve the preceding object at microwave frequencies.
It is a further object of the invention to achieve one or more of the preceding objects while selectively effecting relatively large phase changes at microwave frequencies in a compact length.
It is still a further object of the invention to achieve one or more of the preceding objects with apparatus that is mechanically and electrically reliable and relatively easy and inexpensive to fabricate.
SUMMARY OF THE INVENTION
According to the invention, there is at least one filter section with means for selectively establishing low and high pass states with the cutoff frequencies in each state defining an overlapping hand of operation in both states to produce a phase difference in the overlapping band in the transmission characteristics of the two states. Preferably each section comprises microwave diodes which, when not conducting, may be characterized by an equivalent circuit having substantially an inductance in series with a capacitance to form a series resonant circuit resonant at a frequency above the overlapping or pass band of the system. Typically a section comprises PIN diodes, two of which comprise series arms and one of which comprises a shunt arm and means for biasing the series and shunt arms for conduction and nonconduction during mutually exclusive time intervals.
Numerous other features, objects, and advantages of the invention will become apparent from the following specification when read in connection with the accompanying drawing in which:
BRIEF DESCRIPTION OF THE DRAWING
FIG. 1 is a combined diagrammatic and schematic circuit diagram of a section according to the invention;
FIGS. 2A and 2B are schematic circuit diagrams of equivalent circuits of a section under states A and B, respectively, corresponding to low-pass and high-pass states, respectively;
FIGS. 3A and 3B are schematic circuit diagrams of the equivalent circuits of FIGS. 2A and 2B, respectively;
FIG. 4 is a graphical representation of insertion loss as a function of frequency to illustrate how the operating band is in the overlapping bands of low and high pass states;
FIG. 5 is a representation of the circuits of FIGS. 3A and 3B helpful in analysis for transmission phase shift;
FIGS. 6A and 6B are the circuits of FIGS. 3A and 3B with normalized values indicated for 180° phase shift;
FIG. 7 is a schematic circuit diagram of a section according to the invention offering certain practical advantages; and
FIGS. 8A and 8B are schematic circuit diagrams of the equivalent circuits of FIG. 7.
DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS
With reference now to the drawing, and more particularly FIG. 1 thereof, there is shown a combined pictorial schematic circuit representation of an embodiment of the invention. So as not to obscure the principles of the invention, details of the ground planes, terminals and the like are omitted, these aspects of the structure being apparent to those skilled in the art. A typical section may include an input line 11 and an output line 12 characterized by an impedance Z o , typically 50 ohms. Three diodes, such as PIN diodes, 13, 14 and 15 are connected poled as shown to a junction section 16 which receives a biasing potential through r-f choke 17. The anode of diode 15 is connected to ground through line section 18. The cathodes of diodes 14, 15 and 16 may be connected in series with inductive sections 21, 22 and 23, respectively, if additional series inductance is required for a specific operating frequency range.
When series diodes 14 and 13 are forward-biased, shunt diode 15 is reverse-biased. Conversely, when diode 15 is forward-biased, diodes 14 and 13 are reversed-biased.
The equivalent circuit of a diode, such as a PIN diode in a glass package, typically comprises at zero or reverse bias the series combination of lead inductance L, junction capacitance C j and equivalent series resistance in the reverse-bias state R r . At forward bias the lead inductance L is in series with an equivalent forward-bias sereis resistance R f Neglecting losses, the equivalent circuit of a section in low-pass state A with the series diodes forward-biased and the shunt diode reverse-biased is shown in FIG. 2A and in high-pass state B with the series diodes reverse-biased and the shunt diode forward-biased is shown in FIG. 2B. X L is the inductive reactance of the diode lead inductance in series with an inductive section. The impedance X c is the capacitive reactance of a reverse-biased junction. If the circuit is operated below the series resonance of an arm, X L is less than X c and the circuits of FIGS. 2A and 2B may be represented by the circuits of FIGS. 3A and 3B, respectively. X c ' =│X L - X c │. As the bias state of the diodes are switched, the circuit changes from a low pass filter to a high pass filter. By establishing the parameter values so that the cutoff frequency of the low pass state is above the operating band and that of the high pass state is below the operating band, the phase shifter will exhibit low insertion loss in both states in the operating band as indicated in FIG. 4. However, the low pass filter in state A imparts a phase delay while the high pass filter in state B imparts a phase advance. The result is a low-loss transmission section in the operating band having means for selectively shifting between phase delay and phase advance.
Referring to FIG. 5, there is shown a representation of the arms in the circuit sections of FIGS. 2A and 2B helpful in analyzing the phase delay per section.
Δ Φ = Φ A - Φ B = 2 tan -1 [ 1 - (x p - x s ) 2 + x p 2] /2 (x p - x s )
As an example, if Z o = 50 = X L = X c ' = 50 ohms (X c = 100 ohms), the circuit may be represented on a normalized basis as having a vector impedance + j in the series arms and - j in the shunt arm in state A as shown in FIG. 6A. In state B the series arm impedances are each - j and the shunt arm impedance is + j as shown in FIG. 6B. By inspection in this special case it can be seen that the phase delay in state A is + 90° and - 90° in state B to produce a change of 180° between states A and B with one section. The physical realization in strip line is a unit less than an inch long at microwave frequencies while in microstrip the length would be less than one-half inch. Low values of phase shift may be obtained by changing the values of L and C so that x s is less than unity and x p is greater than unity in the equation for phase delay above.
In an actual embodiment of the invention at a frequency of 2.0 GHz, the loss in state A was only 1.60db and that in state B 3.8 db while providing a phase shift difference of 174° between the two states. That embodiment included 0.6 picofarad glass PIN diodes in the series arms and a 0.4 picofarad PIN diode in the shunt arm.
Referring to FIG. 7, there is shown a schematic circuit diagram of a preferred embodiment of the invention for even further reducing insertion loss. This embodiment includes input and output terminals 31 and 32 and a common or grounded line 33. Each series branch comprises a diode D1 shunted by a capacitor of value C1 in series with an effective inductance of value L1 between a respective one of input and output terminals 32 and junction 34 to which a biasing potential is applied. The shunt network comprises a diode D2 in series with an inductance L2 between junction 34 and grounded line 33, the latter series combination being shunted by a capacitance of value C2. Diodes D1 and D2 are extremely low capacitance diodes having a capacitance typically 0.05 to 0.15 picofarads and function essentially only as switches. Capacitors C1 and C2 are preferably high Q chip capacitors. The shunt arm comprises a parallel LC resonant circuit and is advantageous because it leads to more practical element values. Principles of operation are essentially those described above in connection with the circuit in FIG. 2.
In low pass state A, diodes D1 are effectively short circuits while diode D2 is an open circuit. In high pass state B diode D2 is essentially a short circuit while diodes D1 are open circuits. Specific values of L1 and C1, L2 and C2 are chosen so that the circuit then approximates a high pass T filter. With this circuit up to 180° of phase change can be obtained with less than one db loss.
Referring to FIGS. 8A and 8B, there are shown equivalent circuits of the circuit of FIG. 7 in the A and B states respectively. In the A and B states the series diodes are forward-biased and reverse-biased, respectively, and the shunt diode is reverse-biased and forward-biased, respectively. For determining phase shift, it is convenient to assume that the resistances are negligible and may be neglected. Then the transmission phase shift in the two biased states is given by the following expression. ##SPC1##
where x = ωL 1 and b ≉ω (C 2 + C j2 ) ≉ ω C 2 for the A state;
x ≉ ωL 1 - 1/ωC 1 , and
b = ωC 2 - 1/ωL 2 for the B state for the bias states described in FIG. 8B. It is assumed that the junction capacitance of the diodes is so small that 1/ωC j2 >> ω L 2 ; C j2 << C 2 ; and C jl << C 1 .
In other words: ##SPC2##
where X A ≡ │ ωL 1 /Z o │ and B A ≡ │ ωC 2 /Y o │ in the A state. In the B state it is assumed that ωL 1 < 1/ωC 1 and ωC 2 < 1/ωL 2 , then x and b are negative so that the transmission phase shift ##SPC3##
where X B = (ωL 1 - 1/ωC 1 )/Z o and B B = (ωC 2 - 1/ω L 2 )/Y o
If X A = X B = B A = B B = 1 at midband, then the circuits of FIGS. 8A and 8B correspond to the circuits of FIGS. 6A and 6B described above.
It is convenient to define impedance levels in the A and B states as:
K A = √(2L 1 /C 2) and K B = √(L e /2 C e ) respectively, and cutoff frequencies
f A = (1/2π) √(2/L 1 C 2 ) and f B = (1/2π) √(1/2L e C e ) where for frequencies which satisfy ω 2 < (1/L 1 C 1 ) and ω 2 < (1/L 2 C 2 ), L e = L 2 /(1 - ω 2 L 2 C 2 ) and C e = C 1 /(1 - ω 2 L 1 C 1 ). Using these definitions X B = (1/ωC e Z o ) and B B = (Z o /ωL e ).
In order to have a reasonably well-matched structure, K A and K B should be close to Z o . Choosing K A = √ 2Z o and K B = (Z o /√2), Z o = √(L 1 /C 2) = √L e /C e , or (L 1 /C 2) = (L e /C e ) = Z o 2 . This results in X A = B A and X B = B B . Similarly, a reasonable compromise for setting the low and high pass cutoff is (f A /f o) = (f o /f B) where f o = center frequency of the operating band. This results in
f o = 1/2π √ L 1 C e = 1/2π √ L e C 2 .
Hence, f A /f o = √ 2L e /L e and f B /f o = √ L 1 /2 L e .
Manipulation of the above expressions for f o leads to
L 2 = 1/(2 ω o 2 C 2 ) and C 1 = 1/(2 ω o 2 L 1 ).
Given Z o , f o and the required ΔΦ at midband, the following steps may be followed to determine parameter values. 1. Find the midband value of X A (which is ωw o L 1 /Z o) for the required value of ΔΦ from the following expression.
X A [(3 - X A 2 )/(1 - X A 2 )] = 2 tan ΔΦ/2.
The values of X A for common values of ΔΦ are given in the following table. (Note that to a first approximation 2 tan ΔΦ/2 = 3X A .)
Φ 0 111/4 ° 221/2 ° 45° 90° 180° ωl 1 /z o 0 0.07 0.14 0.26 0.53 1
2. Determine C 2 from the equation.
C 2 = L 1 /Z o 2
3. Determine L 2 and C 1 from the equations.
L 2 = 1/2ω o 2 C 2 and C 1 = 1/2ω o 2 L 1 .
It can be shown that values of ΔΦ within 10 percent of a predetermined nominal value may be achieved over a 20 to 30 percent frequency range. However, greater uniformity of phase shift may be achieved within a frequency range not exactly centered about the nominal center frequency f o . It may be advantageous to plot ΦA - ΦB as a function of frequency experimentally or through computer simulation techniques in connection with selecting an optimum frequency range for a predetermined phase shift.
It can be shown that good VSWR in both the A and B states can be obtained over a relatively wide bandwidth for a phase difference up to 90°. For a phase difference of 180°, the VSWR is good over a bandwidth of 10 to 15 percent of center frequency.
It can also be shown that the insertion loss is roughly constant over a 20 percent bandwidth and less than a decibel for a 90° phase shift, and even smaller for lesser phase shifts.
It can also be shown that a unit operating at C band can readily handle 10 watts of incident r-f power. For Z o = 50 ohms and 200 volt PIN diodes, the maximum allowable r-f power before appreciable nonlinearity occurs is about 800 watts peak. For 400 volt diodes, the allowable peak power is roughly 3 kilowatts.
The invention is thus capable of wide band low-loss performance in an exceptionally compact structure especially suitable for microstrip fabrication.
There has been described a novel digital phase shifter characterized by relatively low insertion loss, relatively large available phase shift in an exceptionally compact physical structure while operating reliably and being relatively inexpensive to fabricate. It is evident that those skilled in the art may now make numerous modifications and uses of and departures from the specific embodiments described herein without departing from the inventive concepts. Consequently, the invention is to be construed as embracing each and every novel feature and novel combination of features present in or possessed by the apparatus and techniques herein disclosed and limited solely by the spirit and scope of the appended claims.
The present invention relates in general to phase shifting and more particularly concerns a novel microwave phase shifter that advantageously employs PIN diodes to provide a reliable digital phase shifter with relatively compact inexpensive reliable apparatus. The invention realizes large phase changes at microwave frequencies in a very compact length.
It is an important object of the invention to provide an improved phase shifter.
It is another object of the invention to achieve the preceding object at microwave frequencies.
It is a further object of the invention to achieve one or more of the preceding objects while selectively effecting relatively large phase changes at microwave frequencies in a compact length.
It is still a further object of the invention to achieve one or more of the preceding objects with apparatus that is mechanically and electrically reliable and relatively easy and inexpensive to fabricate.
SUMMARY OF THE INVENTION
According to the invention, there is at least one filter section with means for selectively establishing low and high pass states with the cutoff frequencies in each state defining an overlapping hand of operation in both states to produce a phase difference in the overlapping band in the transmission characteristics of the two states. Preferably each section comprises microwave diodes which, when not conducting, may be characterized by an equivalent circuit having substantially an inductance in series with a capacitance to form a series resonant circuit resonant at a frequency above the overlapping or pass band of the system. Typically a section comprises PIN diodes, two of which comprise series arms and one of which comprises a shunt arm and means for biasing the series and shunt arms for conduction and nonconduction during mutually exclusive time intervals.
Numerous other features, objects, and advantages of the invention will become apparent from the following specification when read in connection with the accompanying drawing in which:
BRIEF DESCRIPTION OF THE DRAWING
FIG. 1 is a combined diagrammatic and schematic circuit diagram of a section according to the invention;
FIGS. 2A and 2B are schematic circuit diagrams of equivalent circuits of a section under states A and B, respectively, corresponding to low-pass and high-pass states, respectively;
FIGS. 3A and 3B are schematic circuit diagrams of the equivalent circuits of FIGS. 2A and 2B, respectively;
FIG. 4 is a graphical representation of insertion loss as a function of frequency to illustrate how the operating band is in the overlapping bands of low and high pass states;
FIG. 5 is a representation of the circuits of FIGS. 3A and 3B helpful in analysis for transmission phase shift;
FIGS. 6A and 6B are the circuits of FIGS. 3A and 3B with normalized values indicated for 180° phase shift;
FIG. 7 is a schematic circuit diagram of a section according to the invention offering certain practical advantages; and
FIGS. 8A and 8B are schematic circuit diagrams of the equivalent circuits of FIG. 7.
DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS
With reference now to the drawing, and more particularly FIG. 1 thereof, there is shown a combined pictorial schematic circuit representation of an embodiment of the invention. So as not to obscure the principles of the invention, details of the ground planes, terminals and the like are omitted, these aspects of the structure being apparent to those skilled in the art. A typical section may include an input line 11 and an output line 12 characterized by an impedance Z o , typically 50 ohms. Three diodes, such as PIN diodes, 13, 14 and 15 are connected poled as shown to a junction section 16 which receives a biasing potential through r-f choke 17. The anode of diode 15 is connected to ground through line section 18. The cathodes of diodes 14, 15 and 16 may be connected in series with inductive sections 21, 22 and 23, respectively, if additional series inductance is required for a specific operating frequency range.
When series diodes 14 and 13 are forward-biased, shunt diode 15 is reverse-biased. Conversely, when diode 15 is forward-biased, diodes 14 and 13 are reversed-biased.
The equivalent circuit of a diode, such as a PIN diode in a glass package, typically comprises at zero or reverse bias the series combination of lead inductance L, junction capacitance C j and equivalent series resistance in the reverse-bias state R r . At forward bias the lead inductance L is in series with an equivalent forward-bias sereis resistance R f Neglecting losses, the equivalent circuit of a section in low-pass state A with the series diodes forward-biased and the shunt diode reverse-biased is shown in FIG. 2A and in high-pass state B with the series diodes reverse-biased and the shunt diode forward-biased is shown in FIG. 2B. X L is the inductive reactance of the diode lead inductance in series with an inductive section. The impedance X c is the capacitive reactance of a reverse-biased junction. If the circuit is operated below the series resonance of an arm, X L is less than X c and the circuits of FIGS. 2A and 2B may be represented by the circuits of FIGS. 3A and 3B, respectively. X c ' =│X L - X c │. As the bias state of the diodes are switched, the circuit changes from a low pass filter to a high pass filter. By establishing the parameter values so that the cutoff frequency of the low pass state is above the operating band and that of the high pass state is below the operating band, the phase shifter will exhibit low insertion loss in both states in the operating band as indicated in FIG. 4. However, the low pass filter in state A imparts a phase delay while the high pass filter in state B imparts a phase advance. The result is a low-loss transmission section in the operating band having means for selectively shifting between phase delay and phase advance.
Referring to FIG. 5, there is shown a representation of the arms in the circuit sections of FIGS. 2A and 2B helpful in analyzing the phase delay per section.
Δ Φ = Φ A - Φ B = 2 tan -1 [ 1 - (x p - x s ) 2 + x p 2] /2 (x p - x s )
As an example, if Z o = 50 = X L = X c ' = 50 ohms (X c = 100 ohms), the circuit may be represented on a normalized basis as having a vector impedance + j in the series arms and - j in the shunt arm in state A as shown in FIG. 6A. In state B the series arm impedances are each - j and the shunt arm impedance is + j as shown in FIG. 6B. By inspection in this special case it can be seen that the phase delay in state A is + 90° and - 90° in state B to produce a change of 180° between states A and B with one section. The physical realization in strip line is a unit less than an inch long at microwave frequencies while in microstrip the length would be less than one-half inch. Low values of phase shift may be obtained by changing the values of L and C so that x s is less than unity and x p is greater than unity in the equation for phase delay above.
In an actual embodiment of the invention at a frequency of 2.0 GHz, the loss in state A was only 1.60db and that in state B 3.8 db while providing a phase shift difference of 174° between the two states. That embodiment included 0.6 picofarad glass PIN diodes in the series arms and a 0.4 picofarad PIN diode in the shunt arm.
Referring to FIG. 7, there is shown a schematic circuit diagram of a preferred embodiment of the invention for even further reducing insertion loss. This embodiment includes input and output terminals 31 and 32 and a common or grounded line 33. Each series branch comprises a diode D1 shunted by a capacitor of value C1 in series with an effective inductance of value L1 between a respective one of input and output terminals 32 and junction 34 to which a biasing potential is applied. The shunt network comprises a diode D2 in series with an inductance L2 between junction 34 and grounded line 33, the latter series combination being shunted by a capacitance of value C2. Diodes D1 and D2 are extremely low capacitance diodes having a capacitance typically 0.05 to 0.15 picofarads and function essentially only as switches. Capacitors C1 and C2 are preferably high Q chip capacitors. The shunt arm comprises a parallel LC resonant circuit and is advantageous because it leads to more practical element values. Principles of operation are essentially those described above in connection with the circuit in FIG. 2.
In low pass state A, diodes D1 are effectively short circuits while diode D2 is an open circuit. In high pass state B diode D2 is essentially a short circuit while diodes D1 are open circuits. Specific values of L1 and C1, L2 and C2 are chosen so that the circuit then approximates a high pass T filter. With this circuit up to 180° of phase change can be obtained with less than one db loss.
Referring to FIGS. 8A and 8B, there are shown equivalent circuits of the circuit of FIG. 7 in the A and B states respectively. In the A and B states the series diodes are forward-biased and reverse-biased, respectively, and the shunt diode is reverse-biased and forward-biased, respectively. For determining phase shift, it is convenient to assume that the resistances are negligible and may be neglected. Then the transmission phase shift in the two biased states is given by the following expression. ##SPC1##
where x = ωL 1 and b ≉ω (C 2 + C j2 ) ≉ ω C 2 for the A state;
x ≉ ωL 1 - 1/ωC 1 , and
b = ωC 2 - 1/ωL 2 for the B state for the bias states described in FIG. 8B. It is assumed that the junction capacitance of the diodes is so small that 1/ωC j2 >> ω L 2 ; C j2 << C 2 ; and C jl << C 1 .
In other words: ##SPC2##
where X A ≡ │ ωL 1 /Z o │ and B A ≡ │ ωC 2 /Y o │ in the A state. In the B state it is assumed that ωL 1 < 1/ωC 1 and ωC 2 < 1/ωL 2 , then x and b are negative so that the transmission phase shift ##SPC3##
where X B = (ωL 1 - 1/ωC 1 )/Z o and B B = (ωC 2 - 1/ω L 2 )/Y o
If X A = X B = B A = B B = 1 at midband, then the circuits of FIGS. 8A and 8B correspond to the circuits of FIGS. 6A and 6B described above.
It is convenient to define impedance levels in the A and B states as:
K A = √(2L 1 /C 2) and K B = √(L e /2 C e ) respectively, and cutoff frequencies
f A = (1/2π) √(2/L 1 C 2 ) and f B = (1/2π) √(1/2L e C e ) where for frequencies which satisfy ω 2 < (1/L 1 C 1 ) and ω 2 < (1/L 2 C 2 ), L e = L 2 /(1 - ω 2 L 2 C 2 ) and C e = C 1 /(1 - ω 2 L 1 C 1 ). Using these definitions X B = (1/ωC e Z o ) and B B = (Z o /ωL e ).
In order to have a reasonably well-matched structure, K A and K B should be close to Z o . Choosing K A = √ 2Z o and K B = (Z o /√2), Z o = √(L 1 /C 2) = √L e /C e , or (L 1 /C 2) = (L e /C e ) = Z o 2 . This results in X A = B A and X B = B B . Similarly, a reasonable compromise for setting the low and high pass cutoff is (f A /f o) = (f o /f B) where f o = center frequency of the operating band. This results in
f o = 1/2π √ L 1 C e = 1/2π √ L e C 2 .
Hence, f A /f o = √ 2L e /L e and f B /f o = √ L 1 /2 L e .
Manipulation of the above expressions for f o leads to
L 2 = 1/(2 ω o 2 C 2 ) and C 1 = 1/(2 ω o 2 L 1 ).
Given Z o , f o and the required ΔΦ at midband, the following steps may be followed to determine parameter values. 1. Find the midband value of X A (which is ωw o L 1 /Z o) for the required value of ΔΦ from the following expression.
X A [(3 - X A 2 )/(1 - X A 2 )] = 2 tan ΔΦ/2.
The values of X A for common values of ΔΦ are given in the following table. (Note that to a first approximation 2 tan ΔΦ/2 = 3X A .)
Φ 0 111/4 ° 221/2 ° 45° 90° 180° ωl 1 /z o 0 0.07 0.14 0.26 0.53 1
2. Determine C 2 from the equation.
C 2 = L 1 /Z o 2
3. Determine L 2 and C 1 from the equations.
L 2 = 1/2ω o 2 C 2 and C 1 = 1/2ω o 2 L 1 .
It can be shown that values of ΔΦ within 10 percent of a predetermined nominal value may be achieved over a 20 to 30 percent frequency range. However, greater uniformity of phase shift may be achieved within a frequency range not exactly centered about the nominal center frequency f o . It may be advantageous to plot ΦA - ΦB as a function of frequency experimentally or through computer simulation techniques in connection with selecting an optimum frequency range for a predetermined phase shift.
It can be shown that good VSWR in both the A and B states can be obtained over a relatively wide bandwidth for a phase difference up to 90°. For a phase difference of 180°, the VSWR is good over a bandwidth of 10 to 15 percent of center frequency.
It can also be shown that the insertion loss is roughly constant over a 20 percent bandwidth and less than a decibel for a 90° phase shift, and even smaller for lesser phase shifts.
It can also be shown that a unit operating at C band can readily handle 10 watts of incident r-f power. For Z o = 50 ohms and 200 volt PIN diodes, the maximum allowable r-f power before appreciable nonlinearity occurs is about 800 watts peak. For 400 volt diodes, the allowable peak power is roughly 3 kilowatts.
The invention is thus capable of wide band low-loss performance in an exceptionally compact structure especially suitable for microstrip fabrication.
There has been described a novel digital phase shifter characterized by relatively low insertion loss, relatively large available phase shift in an exceptionally compact physical structure while operating reliably and being relatively inexpensive to fabricate. It is evident that those skilled in the art may now make numerous modifications and uses of and departures from the specific embodiments described herein without departing from the inventive concepts. Consequently, the invention is to be construed as embracing each and every novel feature and novel combination of features present in or possessed by the apparatus and techniques herein disclosed and limited solely by the spirit and scope of the appended claims.