BACKGROUND OF THE INVENTION
This invention relates to subsurface antennas, and more particularly to highly efficient buried antenna systems which may be rendered immune to military bombardment.
When an antenna is buried in the ground, it suffers energy losses in several ways. First, the signal energy suffers attenuation through rock. This appears as an exponential loss. Second, there is a "refraction" loss at the interface between the earth and air. Third, the received power density is reduced by the "spread" loss.
In the ground, the reduction in power density and the efficiency of an antenna have lead to the use of the concept of a "modified power gain." The power gain of an antenna in air may be defined as the ratio of the power density at a receiver at a distance R from the transmitting antenna to the power density in the antenna, this ratio being multiplied by 4πR2. For the gain measurement in the ground, the gain in air is multiplied by e2aR where "a" is the attenuation constant. Now, the power density at the receiver is Ke-2aR. Thus, the "modified power gain" Gm may be considered independent of distance in the the rock and expressed as Gm - (Ke-2aR (4πR2) e2aR)/power in antenna.
Generally, the modified power gain as defined above will vary with the type of transmission medium. If the transmission medium is unbounded, homogenous, and isotropic and the receiver is spaced at a large distance from the transmitter, then the linear antennas should preferably be placed parallel to each other for maximum power transfer. This is termed broadside radiation.
It is well known that buried antennas near the surface may transfer and receive energy by three mechanisms. The first mechanism is the generation of a surface wave, which is a substantially vertically polarized wave propagating, along the earth's surface. In the second mechanism, one buried antenna generates a space wave which upon reflection may communicate with another buried antenna in the so-called "up-over-and-down" propagation scheme. The third transmission mechanism is directly "through the rock" in which waves are polarized in directions parallel to the buried antenna. Reference is made in this regard to the Proceedings of the Conference of the British Institute of Electrical Engineers, 8-10 Nov. 1967, at pages 313-317, in an article entitled, "Subsurface Radio Communications," by Tsao and deBettencourt and "Progress in Radio Science" by J. T. deBettencourt in the International Radio Scientific Union (URSI) Berkeley, California, 1967, part I, pages 697-767.
In survivable communications, such as at missile sites, it is anticipated that the requirements to survive in the event of nuclear attack would indicate the use of subsurface antennas for radio communications. Furthermore, it is contemplated that any one or all of the three propagation mechanisms would be used. Needless to say, these three mechanisms are subject to one or more of the aforementioned power or energy losses.
In the prior art, such as U. S. Pat. No. 3,346,864, issued to G. J. Harmon on Oct. 10, 1967, it is taught that an underground antenna may be positioned at an inclination greater than 10° to the horizontal and at a depth greater than 1/10 the wavelength. If the electrical conducting antenna surface is placed in close contact with the subsurface rock, then a space wave is propagated. However, this antenna system produces a space wave component in the direction of interest. Such a system requires surfaces inclined at a specific angle of inclination.
It is accordingly an object of this invention to devise a subsurface antenna useful in transmitting in the three principal propagation mechanisms and capable of being placed in both plane and irregular topography.
In addition to the Harmon reference, another patent relating to underground antennas is U. S. Pat. No. 3,183,510. This patent does not show systems which optimize the surface wave, space wave, and broadside radiation so as to increase power gain of the antenna. The prior art furthermore deals only with antennas 1/2 wavelength long and of the standing wave type.
It is accordingly another object of this invention to devise a geometrically simplified subsurface antenna structure which optimizes antenna gain and is capable of directionality.
SUMMARY OF THE INVENTION
The foregoing objects of this invention are satisfied in an embodiment in which an insulated linear radiating element terminated in its characteristic impedance, that is matched termination, is positioned below the surface for coupling a portion of the surface wave component of an incident electromagnetic wave. A plurality of reactive impedance elements are serially distributed and interconnect portions of the radiating element at periodic intervals along its extent. This permits matching the phase velocity of the induced wave to the phase velocity in the propagation medium, be it air or ground. This matching permits the most effective antenna gain and control of directionality.
In one embodiment, the means for speed matching comprise a plurality of capacitors periodically interposed along a horizontally placed radiating element. In another embodiment, a plurality of inductances are periodically interposed along the extent of a vertically placed radiating element. In both instances, the radiating element has a matched termination. This results in a traveling wave because of the freedom from reflections at the termination. In this regard, it is helpful to consider the radiating element as a transmission line terminated in its characteristic impedance. By using capacitive elements in the horizontal case, the induced wave is speeded up to match the surface wave. In the vertical case, the induced wave is slowed down to match the propagation through the rock.
Either a space wave for the up-over-and-down mode of propagation may be generated or a surface wave generated by a subsurface antenna. However, in the vertical case, because of the depth of attenuation effect in the ground, the lower portion of the vertical antenna is not as effective as the upper portion in contributing to the field in air. Thus, there is a maximum useful length in the vertical extent for these transmission mechanisms.
If the radiating element current propagation constant k2 is equal to B2 - ja2 where B2 is equal to the phase constant and a2 is the attenuation constant, then the useful radiation element length Leff is governed largely by the exponential factor e-a2L, where L is the physical length of the element. Thus, when the a2 L has the value of several nepers, further increase in length L will no longer be useful. However, by proper design of the antenna element, B2 can be made to greatly exceed a2. Then, the useful physical length of the radiating element can be many wavelengths long.
The embodiments of this invention transmit in the three principal propagation modes and, because of directionality, are capable of being placed in both plane and irregular topography. The structure by using serially interposed reactive impedance elements epitomizes geometrical simplicity. Since the phase and distance are related along the antenna propagation axis, the vectoral addition may be controlled to effect directionality.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1a shows a vertically disposed subsurface traveling wave antenna;
FIG. 1b shows a horizontally disposed subsurface traveling wave antenna;
FIG. 1c shows a subsurface communication system.
FIG. 2 shows the interaction between a surface wave and horizontal subsurface antenna;
FIG. 3 shows experimental curves between a bare linear radiating element and a capacitively loaded linear radiating element in a dissipative medium;
FIG. 4 shows the amplitude of current distributions on unloaded and the capacitor loaded bare linear raditating elements at cut-off frequency;
FIG. 5 shows the phase shifts of the current waves on the bare linear radiating elements;
FIGS. 6a and 6b show the amplitude and phase angle of current with and without capacitive loading on an insulated linear radiating element.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
In FIG. 1a, an insulated linear radiating element, such as a type RG58/U cable (outer metallic braid and protective neoprene jacket removed) is vertically disposed at a distance D under the ground. A plurality of inductances 2a interconnect portions of the inner conductor 3 along its extent. The inner conductor 3 is surrounded by an insulating dielectric 4. The surface end of the inner conductor 3 is shown terminated in a matched load 1. The other end of inner conductor 3 is terminated in either a receiver or transmitter 5 and hence to ground 6.
In FIG. 1b, inner conductor 3 has periodically interposed along its extent a plurality of capacitors 2b. One end of the inner conductor is terminated in load 1 with the other end of the inner conductor being terminated in ground 6 through either a receiver or transmitter 5.
If a bare antenna wire, such as inner conductor 3 was placed in intimate contact with the ground, the wavelength of energy λr propagating down the antenna would approximate the wavelength of energy λg propagating in the ground. When inner conductor 3 is insulated by dielectric 4, then the wavelength on the wire λr approximates the wavelength in the dielectric at best if it unloaded.
In FIG. 2, receiver 5 is shown terminating one end of an antenna 3. Load r terminates the other end of inner conductor 3 to ground. The antenna is assumed sufficiently near the surface such that substantial attenuation effects due to depth may be ignored. Taking the propagation of a wave in air and earth as two of the media of interest, the following remarks will illustrate the relevent principles.
A surface wave with a substantial vertically polarized component 10 is assumed to be propagating along the earth-air interface from right to left. Arbitrary points 1e, 2e, 3e, and 4e are arbitrarily selected along the extent of antenna 3. Corresponding points 1, 2, 3, and 4 are taken at the interface. Now, the phase delay B in time over a horizontal distance d is represented, for example, as d43 /λa (2π). λa is the wavelength of the surface wave propagating in air. A portion of the surface wave is refracted into the earth and induces a wave upon the antenna. The phase delays are identical for distance 1 to 1e, 2 to 2e, etc. On the antenna, the wave propagates again from right to left but with a velocity ve. The corresponding phase delay is (2πd5,3e)/λe, where λe is the wavelength on the antenna. If the total delay for the wavelength from point 4 to point 5 (the receiver) is the same regardless of whether the path involves 4, 4e, 3, 3e, etc., then the waves will combine in phase in the receiver. To achieve this, it is necessary that the wavelength in the air λa be the same as the wavelength on the antenna λe. That is, the wavelengths must be equal. Since the velocity va in air is higher than the velocity on an antenna in the earth (va > ve), it is desirable to adjust the phase velocity on the antenna to match that of the surface wave. This is accomplished in the invention by capacitive loading.
This demonstration may be repeated for points 3, 2, and 1. If the phase velocities are matched, then it is possible to have a predetermined vectoral reinforcement at the receiver 5. From this analysis it is clear that, given the use of an insulated wire and further given that the wave velocity on the insulated wire is normally intermediate between the free space velocity in air and the lossy surrounding medium, then the velocity on the insulated wire may be increased if capacitive elements are used and decreased if inductive elements are used.
For surface wave radiation using a traveling wave antenna, the phase constant of the antenna should be nearly equal to the phase constant in the propagation medium. One embodiment of this invention was tested in a salt water model tank to verify the effects of capacitor loading on the characteristics of linear antennas in a dissipative medium. The experiments were conducted in a tank 18 feet in diameter and 3 feet deep. The test frequencies were in the 2-30 megahertz range. The conductivity of the water was maintained at 4 mhos per meter at room temperature. Under these conditions, a 15 inch bare linear radiating element will be electrically long, and the depth of the tank will be much greater than the skip depth.
If the bare antenna is appropriately loaded, the input impedance will be capacitive at frequencies below cut-off and inductive above cut-off. The cut-off frequency is the frequency at which the phase velocity is infinite. That is, the wavelength on the antenna is infinite. The resistance has a minimum value at cut-off. Measurements were made with a bare radiating element suspended below a floating ground plane and driven against it. The capacitor loaded monopole is essentially a series connection of 0.15 microfarad condensers spaced along the monopole length at every 1.7 inches.
In FIG. 3, the impedance versus frequency characteristics of the loaded and unloaded radiating elements are compared. It is observed that resonance occurred at 2 megahertz. Resonance in this context occurs at the cut-off frequency. In order to make true measurements of the attenuation and phase constants, the radiating elements were lengthened to 30 inches. The long radiating elements were placed horizontally in the tank a few inches below the water surface. The ground plane was taken perpendicular to the antenna axis and water surface.
In FIG. 4, for either antenna the current decreases exponentially with distance from the feed point. The corresponding attenuation constant was 5.7 nepers per meter for the unloaded antenna. This was reduced to 3.8 nepers per meter for the capacitor loaded antenna.
In FIG. 5, the phase retardation on the bare unloaded antenna corresponded to a phase constant of 5.5 radians per meter. The phase shift on the capacitor loaded antenna was not measurable. This implies that the wavelength on the antenna due to loading has become infinite.
In FIGS. 6a and 6b the data on the curves shown are measurements taken on an antenna without matched termination. Thus, the current in the antenna exhibits a standing wave pattern having maximum and minimum current amplitudes separated by a distance of a quarter wavelength. These figures clearly demonstrate the loading effects of the propagation constant in the manner previously discussed.
In FIG. 6a on the left hand side a normalized current amplitude I/Ireference, plotted against the antenna length in meters. Phase angle in degrees lag is plotted against antenna length in meters on the right hand side. The applied signal frequency was 21 megahertz.
FIG. 6b shows experimental measurements of the same antenna of three meters length. The antenna is insulated and capacitively loaded with 400 uuf capacitors spaced at 20 centimeters apart. The applied signal frequency was taken at 21 megacycles. The normalized current gain is scaled on the left while phase lag is scaled on the right. Comparison of the figures shows that the current amplitude on the capacitively loaded antenna shows an almost linear relation with length. In contrast, the unloaded antenna is characteristically non-linear. Similar results are apparent when comparing phase. Extrapolating to the case of terminating the antennas in their characteristic impedance, it is clear that the current phase components will add in the capacitor loaded case.
In summary, a subsurface antenna useful in transmitting or receiving a surface wave, space wave, or ground wave energy components has been shown in which the velocity of the wave either radiating from or induced upon the antenna is matched to that of the propagating medium. This matching permits a more efficient energy transfer and further increases directionality.