04/699149

01/05/1971

01/19/1968

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Chicago Musical Instrument Co. (Chicago, IL)

84/668

84/714, 984/351

G10H1/40

84/1.01,1.03,1.24,1.25,1.26

Ray W. E.

I claim

1. A musical-rhythm-producing sound device comprising:

2. A rhythm-producing sound device according to claim 1 further including a chain of flip-flop circuits interposed between said tempo oscillator and said ring counter, and wherein said means connected between said tempo oscillator and said ring counter for changing said ring counter includes means for switching said chain of flip-flops from a divide-by-four condition to a divide-by-three condition.

3. A musical-rhythm-producing rhythm-producing sound device comprising:

4. A rhythm-producing sound device according to claim 3 wherein said gate means also receives output signals from said chain of flip-flops interposed between said tempo oscillator and said ring counter.

5. In a rhythm-producing sound device comprising in combination:

6. A rhythm-producing sound device according to claim 5 wherein said means for changing the rate of operation of said generator means causes said generator means to be changed by an amount substantially proportional to the fraction of the pulse time cycle over which said circuit means shifts the time position of said generator means.

7. A rhythm sound-producing device comprising: a time-point generator for developing a plurality of sequentially generated drive pulses and a plurality of sequentially generated gate pulses, said time-point generator including:

8. An instrument sound generator according to claim 7 wherein said matrix assembly means provides a direct current coupling for said drive pulses and said gate pulses to said keying means.

9. An instrument sound generator according to claim 8 further including means connected between the last stage of said ring counter and the third stage of said ring counter for selectively changing the numerical size of said ring counter from eight to six.

10. An instrument sound generator according to claim 9 further including means connected between the output of the fourth of said flip-flops of said beat divider to the input of the third of said flip-flops of said beat divider to selectively change the count capacity of said third and fourth flip-flops from a divide-by-four count to a divide-by-three count.

11. A rhythm sound-producing device comprising:

12. A rhythm sound-producing device according to claim 11 wherein said clock generator is formed by a tempo oscillator and a flip-flop circuit connected to the output of said tempo oscillator, and means connected from the output of said flip-flop circuit to the input of said tempo oscillator for changing the period of one-half cycle of oscillation from said oscillator.

13. A rhythm sound-producing device according to claim 11 wherein said clock generator is formed by a free-running multivibrator having two symmetrical transistor circuits for producing symmetrical square wave signals, and further including resistor and switch means connected in circuit with one of said transistors for unbalancing the circuit to produce nonsymmetrical square wave pulses.

14. A rhythm sound-producing device according to claim 11 further including means for selectively changing the numerical size of said counting means.

15. A rhythm sound-producing device comprising:

16. A musical-rhythm-producing sound device comprising:

17. A musical-thythm-producing sound device comprising:

18. An instrument sound-generating device comprising:

19. An instrument sound generator according to claim 18 further including circuit means connected between one stage of said ring counter and some other stage thereof for decreasing the number of effective stages in said ring counter.

20. An instrument sound generator according to claim 18 further including means for changing the frequency of the pulse signals produced by said tempo oscillator, and further including means connected between one stage of said ring counter and some other stage thereof for decreasing the number of effective stages in said ring counter.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention is directed generally to an electronic rhythm-producing sound device for automatically generating sounds in rhythmic patterns for use alone or in accompaniment with other musical instruments. Specifically, the invention is directed to a new and improved circuit arrangement for a rhythm-producing sound device.

2. Description of the Prior Art

Heretofore, automatic rhythm devices use electronic clock means which would produce evenly spaced pulses to drive ring or binary counters of a fixed number of stages. The proper pattern of pulses for producing rhythmic patterns were then obtained by gating the proper pulses from the chain each time the counter cycled. A large number of evenly spaced pulses had to be included in the cycle in order to permit all the desired patterns to be obtained. Accordingly, the present invention decreases the number of pulses necessary by providing unevenly spaced pulses when required. In so doing, the gating and logic circuitry required to obtain the desired pulse pattern is greatly simplified. In addition to the simplification resulting from working with a fewer number of pulses and of more closely fitting the pattern to be produced, the present invention also uses the more economical resistor matrices instead of the diode matrices of prior-art devices.

SUMMARY OF THE INVENTION

The device includes a time-point generator, a plurality of direct current coupled matrix patterns, and an instrument generator. The time-point generator comprises a tempo oscillator which generates a series of pulses to control the rate of the rhythm device. The frequency of the tempo oscillator is adjustable. The pulse signal information from the tempo oscillator is delivered to a series of beat dividers which include a feedback path from the fourth stage to the third stage to make the overall scale of the series of dividers either 16 or 12. The tempo oscillator and beat divider together form a clock generator. In the scale of 16 mode of operation, the intermediate beats are evenly spaced throughout the bar of music as sixteenth notes, while in the scale of 12 position they are of odd spacing and occur as triplet notes. The last stage of the binary dividers is connected to an eight-stage ring counter. The ring counter generates eight pulses. The last stage of the ring counter is connected to a second chain of binary dividers which has a total count capacity of eight. The second chain of dividers constitutes a bar counter whose output is decoded and used to selectively control changes in the pattern of music on only the second, fourth and eighth bar of the pattern. The outputs from the tempo oscillator, beat dividers, and ring counter are delivered to pulse amplifiers which reshape the outputs into fast-rise short-duration pulses which occur at different points in time depending on the origin of the pulse.

The pulses derived from the foregoing are delivered to the matrix assembly where they are combined to form subpatterns as required by the different arrangements of music. There is a plurality of patterns or arrangements which can be selected by a selector switch, and each of the patterns is derived from its own subpattern or matrix output. In musical arrangements where more than one instrument sound requires the same drive pattern, the matrix producing that particular subpattern is duplicated so that there are separate but identical drives for each musical instrument sound generator.

The final major component of the rhythm device of the present invention is the musical instrument sound generator which includes keying circuits to operate corresponding multivibrators. The keying circuit includes means to reshape the square wave pulses of the multivibrator into wave forms which approximate those produced by actual musical instruments. These reshaped wave forms are delivered to audio amplifiers for reproduction in a loudspeaker system.

The rhythm device of the present invention is energized by a direct current voltage power supply which produces, for example, 5 and 10 volts for the transistor circuitry and a negative 3.5 volts for the matrix assemblies. The power supply is preferably voltage regulated with zener diodes to maintain the voltage within the prescribed limits while variations from 90 to 130 volts AC may occur at the input line.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram showing generally the circuit arrangement of a rhythm sound-producing device constructed in accordance with principles of this invention;

FIG. 2 is divided into two segments designated FIGS. 2a and 2b and shows a detailed schematic diagram of the circuit arrangement of the time-point generator of FIG. 1;

FIG. 3 is divided into two segments designed FIGS. 3a and 3b and shows a detailed schematic diagram of the circuit arrangement of an instrument sound generator of FIG. 1;

FIGS. 4, 5, 6, 7 and 8 show the arrangement of several different matrices which can be connected between the time-point generator and the instrument sound generator of FIGS. 2 and 3;

FIG. 9 is a simplified block diagram showing the general circuit arrangement of the beat divider circuit incorporated in the time-point generator of FIG. 1;

FIG. 10 is a series of wave forms illustrating the operation of the beat divider circuit of FIG. 9;

FIG. 11 is a schematic diagram of an automatic follow circuit arrangement which may be used in conjunction with the rhythm sound device of the present invention;

FIG. 12 is a schematic diagram of an alternate arrangement of an automatic follow circuit which may be used in conjunction with the rhythm sound-producing device of the present invention;

FIG. 13 illustrates a wave form generated by the automatic follow circuits of FIGS. 11 and 12;

FIG. 14 is an alternate arrangement of a clock generator circuit which can be used in accordance with the principles of this invention; and

FIG. 15 is still another alternate arrangement of a clock generator which can be used in accordance with the principles of this invention.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Seen in FIG. 1 is a rhythm sound-producing device constructed according to this invention and includes a time-point generator 10 which produces a plurality of sequential spaced-apart pulses which may correspond to the sequential spacing of a line or bar of written music. The time-point generator 10 includes a tempo oscillator 11 the frequency of which is selectively variable by a potentiometer 12. The potentiometer 12 is connected to a voltage divider network 13 comprising a resistor 14 and a potentiometer 16. The potentiometer 16 is used to calibrate the voltage applied to the tempo oscillator thereby calibrating the range of frequency of oscillation.

The output of the tempo oscillator 11 is delivered to a beat divider circuit 17 and to one of a plurality of pulse amplifiers designated generally by reference numeral 18. The beat divider network 17 is formed by a plurality of flip-flop circuits. A switch 19 is connected to the beat divider network 17 to selectively change the voltage applied to a portion of the beat divider network thereby enabling the network 17 to divide by four or divide by three as desired. The output of the beat divider network 17 is delivered to certain ones of the plurality of pulse amplifiers 18 and to the input of a ring counter 20.

The tempo oscillator 11 together with the beat divider circuit 17 form a clock generator which controls the function of the rhythm device. A clock generator so formed in accordance with this invention can selectively produce evenly or unevenly spaced pulse signals to provide a plurality of different kinds of pulse patterns corresponding to the different kinds of musical arrangements.

Also, according to the present invention, the ring counter 20 is a direct-coupled self-biasing circuit which, in the preferred embodiment, has eight stages. It will be understood that the ring counter of the present invention may have more or less than eight stages.

The ring counter 20 is biased such that seven of the eight stages are conducting in the saturated mode of operation and one stage is nonconducting at all times. Pulses from the beat divider network 17 will cause the nonconducting stage to become conductive and the next preceding stage to become conductive.

The eight stages of the ring counter are divided into two groups, one group containing the odd-number stages and the other group containing the even-number stages.

Connected to the ring counter 20 is a time switch 21 which enables the ring counter to generate a 4/4 time or a 3/4 time depending upon the type of music being reproduced by the rhythm sound generator. By way of example, the ring counter 20 is an eight-stage counter wherein the output of the last stage is fed back to the input of the first stage thereby allowing the ninth pulse from the beat divider to restart the count by eight cycle. With the time switch 21 in the open position, as shown on the drawing, the ring counter will produce eight pulses in sequence. However, when the time switch 21 is in the closed position, the first two stages of the ring counter are bypassed, thereby converting the ring counter into a count by six circuit. This feature enables the ring counter 20 to generate both 4/4 and 3/4 time counts for the various types of music being reproduced.

The plurality of outputs of the ring counter 20 are connected to still another group of pulse amplifiers 18 and to a bar counter 22. The bar counter 22 is formed by a chain of three binary stages or flip-flops having a total binary-counting capacity of eight which is a musically correct bar structure. Each time the ring counter 20 produces the eight pulse of the count by eight sequence of operation it generates an output pulse at the input of the bar counter 20. The output of the bar counter is decoded by a series of logic circuits, which will be described hereinbelow, and applied to a matrix assembly 23. Also connected to the bar counter 20 is a reset switch 24 which resets all of the flip-flop circuits of the bar counter to the zero count.

Also connected to the matrix assembly 23 are the outputs of the pulse amplifiers 18. The matrix assembly 23 is constructed primarily of resistors which are direct current coupled to the pulse amplifiers and bar counter.

The output of the matrix assembly 23 is connected to a selector switch assembly 26 and therefrom to an instrument sound generator 27. The selector switch 26 connects the proper matrices within the matrix assembly 23 to the respective ones of a plurality of instrument driver circuits 28. The instrument drivers 28 are connected to one or more of a plurality of instrument generator circuits 29. The output of the instrument generators are then applied to mixing amplifiers designated generally by reference numeral 30.

The output of the mixing amplifiers is connected to an audio amplifier (not shown) through a coupling circuit comprising a voltage divider network which is formed by resistor 31 and a potentiometer 32. A capacitor 33 has one end thereof connected to the movable contactor of the potentiometer 32 and the other end thereof connected to one end of the resistance element of the potentiometer 32. The audio signal passing through the capacitor 33 and the movable contactor of the potentiometer 32 is delivered to a shielded cable 34 through a resistor 36 and a coupling capacitor 37. The output of the mixing amplifiers 30 is connected to the coupling circuit through a shielded cable 38. Connected to the junction of the movable contact of the potentiometer 32 and the capacitor 33 is a switch 39 which is connected to ground potential. When the switch 39 is closed, the audio signal from the mixing amplifiers 30 is grounded to mute the output of the rhythm sound device.

FIG. 2 is a detailed schematic diagram of the time-point generator 10 of FIG. 1 and is divided into two segments FIG. 2a and FIG. 2b. The tempo oscillator is designated generally by reference numeral 40 and includes a unijunction transistor 41. The transistor 41 has the first base electrode thereof connected to ground potential and the second base electrode connected to a positive 10 volt direct current power source, (not shown), through a resistor 42. A charging capacitor 43 is connected between the emitter electrode of the unijunction transistor 41 and ground potential and receives charging current through a resistor 44 and a potentiometer 46. The potentiometer 46 is connected to a voltage divider network 47 comprising a potentiometer 48 and a resistor 49.

The tempo oscillator is operated as a free-running unijunction relaxation oscillator. The charging capacitor 43 receives charging current through the resistor 44 and the potentiometer 46. When the charging capacitor 43 is charged sufficiently to cause an appreciable current flow to pass between the emitter and base 1 of the unijunction transistor, the capacitor 43 is quickly discharged through the emitter base junction of the transistor. The emitter current flowing through the transistor 41 greatly reduces the interbase resistance of the transistor to cause an increase in current flow between base 1 and base 2. The increase in current flow between the base electrodes of the transistor cause a voltage drop to be developed across the load resistor 42 thereby producing a pulse at the output of the tempo oscillator. When the voltage across the charging capacitor 43 decreases sufficiently to increase the emitter base 1 resistance thereof, the emitter stops conducting and the process repeats itself. This action produces a sawtooth wave form at the emitter junction of the transistor 41 and a series of sharp negative-going pulses at the second base electrode of the transistor and across the load resistor 42. The pulses produced by the tempo oscillator are the basic timing signals for the rhythm sound-producing device of the present invention. By varying the resistance of the potentiometer 46, the charging time of the charging capacitor 43 is changed to change the operating frequency of the oscillator. The potentiometer 48 of the voltage divider network 47 is used to calibrate the tempo oscillator when the tempo control potentiometer 46 is set at its minimum resistance value. Therefore, the maximum frequency of the tempo oscillator 40 is preadjusted by the calibration resistor 48.

Pulse signal information is taken from the emitter junction of the transistor 41 and delivered to a pulse amplifier stage 50. The pulse amplifier stage 50 comprises one of the pulse amplifiers 18, shown in FIG. 1. A transistor 51 has the emitter electrode thereof connected to ground potential and the base electrode thereof connected to a positive voltage source through a biasing resistor 52. The collector electrode of transistor 51 is connected to the voltage source through a load resistor 53. The pulse signal information from the emitter of transistor 41 is delivered to the base of transistor 51 through a coupling network comprising resistor 54 and capacitor 56. The sawtooth wave form generated at the emitter of unijunction transistor 41 is then transformed into a pulse signal and applied to the output terminal T ₁ of the pulse amplifier stage 50.

All of the pulse amplifiers used in the time-point generator of the present invention are constructed in substantially the same manner as that of the pulse amplifier 50 and it will be understood that each pulse amplifier operates in precisely the same manner.

In operation, when no pulse signal information is applied to the base electrode of the transistor 51 the transistor is biased through resistor 52 in the saturated mode of operation. That is, sufficient base current is supplied through resistor 52 to allow current flow to pass between the emitter collector junction of the transistor limited only by the collector load resistor 53.

When a negative-going pulse is applied to the base electrode of transistor 51 through the coupling capacitor 56, the transistor 51 is rendered nonconductive thereby decreasing the collector current and increasing the collector voltage to produce an output pulse at terminal T ₁ . The coupling capacitor 56 quickly charges through resistor 54 and renders the transistor 51 conductive in the saturated mode of operation. It will be noted that only negative-going pulses applied to the base electrode of transistor 51 will cause pulse outputs to be developed thereby. Positive-going pulses would only tend to cause the transistor 51 to conduct harder thereby having no effect at the output terminal T ₁ . The coupling resistor 54 and coupling capacitor 56 are selected to form a differentiating network which is responsive only to rapid changes in the pulse signal information applied therethrough.

The pulse signal information developed across the load resistor 42 of the tempo oscillator circuit 40 is delivered to the first of a series of divider networks or flip-flop circuits 57, 58, 59, and 60. Each of the flip-flop circuits 57--60 is constructed in substantially the same manner, and, for purposes of simplicity, only the flip-flop circuit 58 will be discussed in detail.

The flip-flop circuit 58 includes a pair of transistors 61 and 62 having their emitter electrodes connected to ground potential. The collector electrode of transistor 61 is connected to the base electrode of transistor 62 through a coupling network comprising capacitor 63 and resistor 64. Similarly, the collector electrode of transistor 62 is connected to the base electrode of transistor 61 through a coupling network comprising capacitor 66 and resistor 67. The collector of transistor 61 is connected to a positive voltage source through a load resistor 68 and a voltage-dropping resistor 69. The collector electrode of transistor 62 is connected to the positive voltage source through a load resistor 70 and the resistor 69.

The flip-flop stage 58, as well as flip-flops 57, 59, and 60, is basically a two-stage, direct-coupled amplifier with positive feedback. Initially, either transistor 61 or transistor 62 must be in the saturated mode of operation. For example, if transistor 62 is in the saturated mode of operation, the collector voltage is very near ground potential. This low voltage is coupled to the base electrode of transistor 61 and is sufficient to hold the transistor 61 in the cutoff condition. Therefore, the voltage at the collector of transistor 61 is, for example, approximately 5 volts and this voltage is coupled to the base electrode of transistor 62 through resistor 64. The high positive potential applied to the base of transistor 62 is sufficient to maintain the transistor in the saturated mode of operation. Input signals applied to the flip-flop stage are differentiated by the input coupling capacitor 72 and the resistor 69 to produce a negative-going pulse at the junction of resistors 68 and 70 and 69. The negative-going pulse is coupled through the collector resistors 68 and 70 and the base resistors 64 and 67. The base of transistor 61 and the collector of transistor 62 are not affected by the negative-going pulse. However, the collector of transistor 61 and the base of transistor 62 are driven slightly negative. This action starts to turn off transistor 62 and turn on transistor 61. Once this action is initiated, the positive feedback in the circuit insures that the transistor 62 will be rendered nonconductive and transistor 61 rendered conductive in the saturated mode of operation. Each time a negative-going pulse is applied to the flip-flop 58 the flip-flop changes its state of conduction. Therefore, two pulses applied to the input of the flip-flop stage will produce a single pulse at the output thereof. Each stage 57--60 operates in exactly the same manner as described above such that the total division accomplished by the stages between the tempo oscillator and the output of stage 60 is 16.

Pulse signal information from the tempo oscillator 40 is delivered to flip-flop stage 57 through a coupling capacitor 71, and the output of flip-flop stage 57 is delivered to the input of flip-flop stage 58 through a coupling capacitor 72. Similarly, the output of flip-flop stage 58 is coupled to the input of stage 59 through a coupling capacitor 73 and the output of stage 59 is coupled to the input of stage 60 through a coupling capacitor 74. It will be understood that the coupling capacitors 71--74 together with their associated voltage-dropping resistor, resistor 69 of stage 58, form a differentiating circuit to produce pulse inputs to each of the flip-flop stages.

The output of flip-flop stage 58 is coupled to the input of stage 59 and to a pulse amplifier circuit 76 through a coupling capacitor 77 and a resistor 78. The pulse amplifier circuit 76 operates in substantially the same manner as that of the pulse amplifier circuit 50. A negative-going pulse at the output of flip-flop stage 58 will render the transistor of the pulse amplifier 76 nonconductive to produce a positive-going pulse at the output terminal T ₂ thereof. Similarly, the output of flip-flop stage 59 is coupled to the input of stage 60 and to a pulse amplifier stage 79 to produce a pulse at the output terminal T ₃ of the pulse amplifier. Also, the output of flip-flop stage 60 is coupled to a pulse amplifier stage 82 through a coupling capacitor 83 and a resistor 84 to produce a pulse at the output terminal T ₄ of the pulse amplifier 82. The flip-flop stage 60 has a second output terminal thereof connected to a pulse amplifier stage 86 through a coupling capacitor 87 and a resistor 88, and to a feedback amplifier circuit 89. The feedback amplifier 89 is connected to the second output of flip-flop stage 60 through a resistor 90, and therefore is direct current coupled to the flip-flop stage 60. The output of the feedback amplifier 89 is connected to the input of one of the transistors of the flip-flop stage 59 through a coupling capacitor 91 and a resistor 92.

The feedback amplifier 89 includes a transistor 93 which has the base electrode thereof connected to a positive voltage source through a biasing resistor 94. Also connected to the positive voltage source is the collector electrode of transistor 93 through a load resistor 96. The base of transistor 93 is connected to the triplet switch 19, shown in FIG. 1, through a resistor 97.

In operation, when the triplet switch 19 is in the closed position, a positive bias voltage is applied through the resistor 97 to the base of transistor 93. The transistor 93 is held in the saturated mode of operation due to the bias supplied through the resistor 97 and through the resistor 94. Therefore, a pulse signal supplied to the base of transistor 93 from the second output of the flip-flop stage 60 has no effect on the amplifier stage and no output signal will be produced at the collector junction thereof. Therefore, the last two flip-flop stages 59 and 60 operate in their normal mode to divide by four. On the other hand, when the switch 19, of FIG. 1 is in the open position, the bias supplied to the base of transistor 93 through resistor 97 is removed. The bias for transistor 93 is now derived from the coupling resistor 90 and the biasing resistor 94. It will be noted that the emitter of transistor 93 is connected to a positive voltage source which is less than the positive voltage source connected to the collector thereof. For example, the emitter is preferably connected to a positive 5 volt source. Therefore, the transistor 93 will conduct whenever the input through coupling resistor 90 is equal to or greater than the positive 5 volts supplied to the emitter of transistor 93. It can be seen that the output of the feedback amplifier 89 will follow the second output of the flip-flop stage 60. Therefore, every third input pulse to the flip-flop stage 59 is fed back from the second output of the flip-flop stage 60 to the input of flip-flop stage 59 thereby resetting the stage 59. This action reduces the total count capacity of the last two flip-flops 59 and 60 from four counts to three counts. The output of the last flip-flop stage 60 is then nonsymmetrical in a ratio of 2:1 or divided at third intervals rather than in halves.

According to the present invention, each of the pulse amplifiers as well as the feedback amplifier 89 are operated in the saturated mode of operation and produce output pulses only when the transistor of each stage is rendered nonconductive. This feature greatly simplifies the circuit arrangement of the rhythm device of the present invention and reduces the possibility of pulses produced by one stage affecting the operation of the other stages in the circuit.

The pulses produced at the output of flip-flop stage 60, are applied to circuit points 98 and 99 which, in turn, are connected to circuit points 98a and 99a which are signal input terminals of an eight-stage ring counter. Although the present invention shows an eight-stage ring counter it will be understood that a ring counter having any number of stages may be used in accordance with the principles of this invention.

According to this invention a single transistor is used in each stage of the ring counter. The ring counter comprises transistors 100, 101, 102, 103, 104, 105, 106, and 107. The emitter electrode of each transistor is connected to ground potential while the collector electrode thereof is connected to a positive voltage source through a corresponding load resistor 108, 109, 110, 111, 112, 113, 114, and 115 respectively. It will be noted that each stage of the ring counter is constructed in substantially the same manner.

The circuit point 98a from the output of the beat divider network 60, is connected to the base electrodes of transistors 100, 102, 104, and 106 through corresponding resistors 116, 117, 118, and 119 respectively. Similarly, the circuit point 99a is connected to the base electrode of transistor 101, 103, 105, and 107 through corresponding resistors 120, 121, 122, and 123 respectively.

The collector electrode of transistor 100 is coupled to the base electrode of transistor 101 through a resistor 124 and a capacitor 126, while the collector electrode of transistor 101 is coupled to the base electrode of transistor 102 through a resistor 127 and a capacitor 128. It will be noted that the outputs of the remaining transistors 102--107 are coupled to the corresponding down-stage transistors through resistive capacitive networks comprising resistors 129, 131, 133, 136, 138 and 140 and capacitors 130, 132, 134, 137, 139, and 141 respectively.

The collector of transistor 100 is direct current coupled to the base electrode of transistors 102, 104, and 106 through resistors 142, 143, and 144 respectively. The collector of transistor 101 is direct current coupled to the base electrode of transistors 103, 105 and 107 through resistors 146, 147, and 148 respectively. The collector of transistor 102 is direct current coupled to the base electrode of transistors 100, 104, and 106 through resistors 149, 150, and 151 respectively. The collector of transistor 103 is direct current coupled to the base electrode of transistors 101, 105, and 107 through resistors 152, 153, and 154 respectively. The collector electrode of transistor 104 is direct current coupled to the base electrode of transistors 100, 102, and 106 through resistors 157, 156, and 158 respectively. The collector electrode of transistor 105 is direct current coupled to the base electrode of transistors 101, 103, and 107 through resistors 160, 159, and 161 respectively. Similarly, the collector electrode of transistor 106 is direct current coupled to the base electrode of transistors 100, 102, and 104 through resistors 164, 163, and 162 respectively, and the collector electrode of transistors 107 is direct current coupled to the base electrode of transistors 101, 103 and 105 through resistors 168, 167, and 166 respectively. It will be noted that the output of the last transistor 107 is coupled back to the input of the first transistor 100 through the resistor 140 and capacitor 141.

In operation, the biasing of each of the transistors 100--107 arranged such that seven of the transistors are operated in the saturated mode of operation while only one of the transistors is nonconductive. The transistors 100, 102, 104, and 106 form one group of count circuits while transistors 101, 103, 105, and 107 form a second group of count circuits. In each group of transistors the base electrode of each transistor is connected to the collector electrodes of the other three transistors in the group. The interconnecting resistors and the collector load resistor of each stage in the group are sized such that any three stages of the group will be held in the saturated mode of operation if the remaining stage is in the cutoff condition. The transistor which is in the cutoff mode of operation will remain cut off as long as the collector voltage of the remaining three saturated transistors remains below the voltage necessary to bias the transistor on.

Each base electrode of transistors 100--107 is coupled to the circuit points 98a and 99a to receive drive pulses therefrom. The drive pulses are developed at the output of the flip-flop stage 60 of the beat dividers. When the second stage of the flip-flop 60 is in the saturated mode of operation, substantially no drive pulse is delivered to the nonconductive transistor of the group of transistors 100, 102, 104 and 106. On the other hand, when the second stage of flip-flop circuit 60 is rendered nonconductive, a positive pulse is applied to the circuit point 98a to drive the nonconductive transistor of the group 100, 102, 104, and 106 into the saturated mode of operation thereby placing all of the transistors in the group in the saturated mode. Therefore, the transistor being rendered conductive produces an output pulse through the resistor capacitor coupling network between each of the transistor stages and the following stage, which is one of the transistors of the second group, is rendered conductive. That is, when the voltage on the circuit points 98a and 99a causes a previously cutoff transistor to be rendered conductive, the drop in collector voltage applied through the resistor capacitor coupling network to the base of the succeeding saturated transistor will cause this transistor to be cut off. As the voltage on the circuit point 99a is removed due to the saturated condition of the first stage of flip-flop circuit 60 the stage which is rendered nonconductive remains nonconductive and holds the other transistors in the group in the saturated mode of operation.

The nonconductive mode of operation is then shifted from one transistor to another and from one group to another as the output of flip-flop circuit 60 changes state. That is, the stage in each group of transistors of the ring counter becomes nonconductive in turn in response to the successive output pulses of the flip-flop circuit 60. During the time each of the transistors is nonconductive, the voltage on the collector electrode thereof goes positive while the voltage developed at the base electrode thereof goes negative. Therefore, either of the voltages developed at each of the transistor stages of the ring counter can be used as an output pulse.

In each of the resistor capacitor coupling networks between the successive transistors of the ring counter the resistor functions as a voltage divider with the resistance networks connected to the base of the preceding stage such that the voltage swing at the base is a fraction of the voltage swing of the collector. This allows the collector voltage swing to be several times the maximum base emitter reverse voltage rating which is quite small in some transistors. This circuit arrangement allows the voltage at the collector electrode to rise sharply when the transistor is cut off. This action drives the other transistors in the group more quickly into the saturated mode of operation and reduces the possibility of spurious operation.

It will be understood that the number of stages in the ring counter could be increased or decreased by changing the number of stages in each group. Also, it will be understood that the ring counter could be arranged such that all but one of the transistors is operated in the cutoff mode while only one of the transistors is in the saturated mode.

The output of transistor 100 is connected to a terminal T ₆ and to a pulse amplifier 170 which, in turn, is connected to a terminal T ₇ . The transistor 101 is connected to a terminal T ₈ and to a pulse amplifier 171 which, in turn, is connected to a terminal T ₉ . The transistor 102 is connected to a terminal T ₁₀ and to a pulse amplifier 172 which, in turn, is connected to a terminal T ₁₁ . The transistor 103 is connected to a terminal T ₁₂ and to a pulse amplifier 173 which, in turn, is connected to a terminal T ₁₃ . Similarly, transistors 104, 105, 106, and 107 are connected to terminals T ₁₄ , T ₁₆ , T ₁₈ , and T ₂₀ and to pulse amplifiers 174, 176, 177, and 178 respectively. The pulse amplifiers 174, 176, 177, and 178 are connected to output terminals T ₁₅ , T ₁₇ , T ₁₉ , and T ₂₁ respectively. At each of the output terminals represented by a square there is produced a gating pulse, while at each of the output terminals represented by a circle there is produced a pulse signal output. The terminals T ₆ --T ₂₁ are arranged for direct current coupling to one or more matrices of the matrix assembly 23, shown in FIG. 1. Also, it will be understood that each of the pulse amplifiers 170, 171, 172, 173, 174, 176, 177, and 178 are constructed and operate in substantially the same manner as the pulse amplifier 50 connected to the tempo oscillator 40.

According to the present invention, the output of the last transistor 107 of the ring counter is connected to the base electrode of transistor 102 through a line 179, a capacitor 180, the switch 21, shown in FIG. 1, and a resistor 181. When the switch 21 is in the closed position, connecting resistor 181 to capacitor 180, the feedback of transistor 107 is coupled to transistor 102 rather than to transistor 100 thereby decreasing the numerical size of the ring counter from eight stages to six stages. This feature enables the time-point generator of the present invention to produce both 4/4 time pulses and 3/4 time pulses necessary for musical arrangements by using the same electronic circuitry.

The output of the last transistor 107 of the ring counter is also coupled to the input of a series of flip-flop circuits 182, 183, and 184. The flip-flop circuits 182--184 are constructed in substantially the same manner as flip-flop circuit 58 and operate similar thereto. The first stage of flip-flop circuit 182 is direct current coupled to a terminal T ₂₂ and to the base of two transistors 186 and 187 through resistors 188 and 189 respectively. The transistors 186 and 187 form NAND gate circuits. The output of the second stage of the flip-flop circuit 182 is connected to a terminal T ₂₃ .

The first stage of the flip-flop circuit 183 is direct current coupled to transistor 186 through a resistor 190 and to transistor 187 through a resistor 191. The second stage of the flip-flop circuit 183 is capacitively coupled to the input of flip-flop circuit 184. The first stage of flip-flop circuit 184 is direct current coupled to transistor 187 through a resistor 192.

The output of transistor 187 is direct current coupled to the base electrode of a transistor 193 through a resistor 194. The transistors 186, 187, and 193 are connected to a positive voltage source through load resistors 196, 197, and 198 respectively. The pulse or gate voltage developed at the load resistors is applied to terminals T ₂₄ , T ₂₅ , and T ₂₆ . The output pulse developed at terminal T ₂₆ will be opposite that developed at terminal T ₂₅ due to the circuit arrangement of transistors 187 and 193. When transistor 187 is in the saturated mode of operation, transistor 193 is cut off thereby producing a positive pulse at terminal T ₂₆ . On the other hand, when transistor 187 is cut off, bias is applied to transistor 193 sufficient to cause the transistor to be saturated and thereby applying ground potential or negative gate pulse to terminal T ₂₆ .

The output of the second stage of the flip-flop circuit 182 and the output of the first stages of flip-flop circuits 183 and 184 are connected to a switching circuit 199 through diodes 200, 201, and 202. The switching circuit 199 includes a transistor 203 which has its emitter electrode connected to ground potential and the base electrode thereof connected to a positive voltage source through a resistor 204. The collector electrode of transistor 203 is connected to the positive voltage source through a load resistor 206. A capacitor 207 is connected across the bias resistor 204. The base electrode of transistor 203 is connected to the reset switch 24, of FIG. 1, through a resistor 208.

When the switch 24 is closed, resistors 204 and 208 form a voltage divider network such that the voltage at the base of transistor 203 is reduced thereby rendering the transistor substantially nonconductive. With the transistor 203 in the nonconductive mode of operation the diodes 200, 201, and 202 are reversed biased thereby allowing the flip-flop stages 182, 183, and 184 to operate in their normal mode of operation. However, when switch 24 is opened, the transistor 203 is rendered conductive in the saturated mode of operation thereby placing ground potential at the cathodes of each of the diodes 200--202. This action will reset the flip-flop stages 182--184 which form the bar counters.

The bar counter formed by the three flip-flop stages 182--184 has a total count capacity of eight which is a musically correct bar structure. Each time the ring counter produces an output pulse at the last stage thereof it triggers the first flip-flop stage 182 of the bar counter. Flip-flop circuits 183 and 184 follow the first stage 182 and accumulate a total of eight different states of operation. In this way the bar counter keeps track of the number of cycles through which the ring counter passes up to eight cycles and then repeats the operation. The output of the bar counter is decoded by the NAND gates connected thereto to produce positive gate pulses during every second, fourth, and eighth bar. These gates are used to effect electrical changes in the matrix assembly 23, of FIG. 1, to introduce pattern changes during every second, fourth and eighth bar.

Since a musical flourish usually occurs during the eighth bar the bar counter is reset to the seventh bar position. This is accomplished by pulling the appropriate collectors of the flip-flop circuits 182, 183, and 184 down through diodes 200, 201, and 202 to force the bar counter to assume the seventh bar position. The reset drive for the flip-flop circuits 182, 183, and 184 is provided by the saturated mode of operation of the switching circuit 199. The resistor 204 maintains the transistor 203 in the saturated condition thereby applying approximately ground potential to the cathodes of the diodes 200, 201, and 202. The diodes 200, 201, and 202 are forward biased and conduct rather than allow the level at the collectors of the transistors of the flip-flops 182--184 to rise so that the bar counter is held in this state. When the switch 24 is closed, ground potential is applied to the base of transistor 203 through resistor 208 thereby rendering transistor 203 nonconductive and raising the voltage developed at the collector thereof. This action reverse biases the diodes 200, 201 and 202 and allows the bar counter to proceed in the normal fashion.

A transistor 210 is connected to the last transistor 107 of the ring counter through a resistor 211. The transistor 210 is connected to a transistor 212 and to a lamp 213. The transistor 210 and 212 and lamp 213 form an indicating circuit which produces a visual light pulse during predetermined time intervals. The light pulse is used as a downbeat indicator for the person who is using the rhythm sound device for accompaniment. Transistors 210 and 212 are arranged in a Darlington configuration to develop a high gain.

FIG. 3 is divided into two portions of 3a and 3b. FIG. 3 is a detailed schematic diagram of the instrument sound generator 27 of FIG. 1. The instrument sound generator of FIG. 3 includes a plurality of free-running multivibrator circuits 220, 221, 222, 223, 224, 225, and 226. The multivibrator 220 is connected to a drive circuit 227 through a diode 228. The multivibrator 221 is connected to a drive circuit 229 through a diode 230, and the multivibrator 222 is also connected to the drive circuit 229 through a diode 231. The multivibrator 223 is connected to a drive circuit 232 through a diode 233. Also connected to the drive circuit 232 is the output of the multivibrator circuit 224. The multivibrator circuit 225 is connected to a drive circuit 236 through a diode 237. Similarly, the output of the multivibrator 226 is connected to a drive circuit 238 through a diode 239.

The output of the multivibrator 220 is delivered to an audio amplifier transistor 240 through a diode 241 and a wave-shaping network 242. The output of the multivibrator 221 is delivered to an audio amplifier transistor 243 through a diode 244 and a wave-shaping network 246. The output of the multivibrator 222 is also applied to the input of the audio amplifier transistor 243 through a diode 247 and a resistor 248. Similarly, the output of the multivibrator 223 is applied to the input of the audio amplifier transistor 243 through a diode 249 and a resistor 250. Also applied to the input of transistor 243 is the output of multivibrator 224 through a diode 251 and a resistor 252.

The output of multivibrator 225 is applied to an audio amplifier transistor 253 through a diode 254 and a pair of resistors 256 and 257 which are center tapped by a capacitor 258. The resistors 256 and 257 and capacitor 258 form a wave-shaping network for the multivibrator 225. The output of the multivibrator 226 is applied to the input of the audio amplifier transistor 253 through a diode 259 and a wave-shaping network 260.

The output of the audio amplifier transistor 253 is direct current coupled to the input of a transistor amplifier stage 261 which in turn, is capacitive coupled to an audio output circuit 262. It will be noted that the output of transistor amplifiers 240 and 243 also is capacitive coupled to the audio output stage 262. The output of the instrument sound generator may include a level control comprising potentiometer 263 and a volume control comprising potentiometer 264. Also, it will be noted that the audio amplifier transistors receive more than one signal from the multivibrator circuits and therefore the amplifiers are also mixing stages to produce in the audio output circuit 262 a composite audio signal.

The instrument sound generator also includes a noise generator circuit 266 which is direct current coupled to the input of the audio amplifier transistor 240 through a plurality of sets of gating diodes 267, 268, 269, and 270. The noise generator produces audio signals of erratic frequencies but which signals complement the quality of the sound generator so as to more accurately simulate actual musical instruments. The noise generator includes a zener diode and a pair of direct current coupled amplifiers. The last stage of the two-stage amplifier has a resistor 266a connected between the collector of the transistor and ground potential. The resistor 266a is equal in value to a resistor 266b which is the collector load resistor of the stage. This circuit arrangement limits the maximum output voltage of the amplifier to one-half of the supply voltage. This feature keeps the amplitude of the noise generator equal to the output of the other signal sources and also keeps the output amplitude constant so that the noise keyers, the sets of diodes 267, 268, 269 and 270, do not feed noise signal information therethrough due to the peaks of the noise signal.

The noise generator 266 is rendered operative by a drive circuit 271, shown on FIGS. 3a and 3b, or by drive circuit 272 shown on FIGS. 3a and 3b. The drive circuits 271 and 272 include a plurality of transistor stages which may be connected to the matrix assembly through the selector switch 26, of FIG. 1. The drive circuit 229 interacts with the drive circuit 272 such that the noise generator 266 may be rendered operative during any one of a plurality of time points generated by the time-point generator of FIG. 2.

According to the present invention, the multivibrators 220--226 are direct current coupled to the inputs of the audio amplifier transistors through gating diode circuits. That is, by way of example, the output of the multivibrator 226 is direct current coupled to the base of audio amplifier transistor 253 through the gating diodes 239 and 259. Also, the drive circuits may contain one or more drive transistors to cause the output of each of the multivibrator circuits to be applied to the audio amplifier transistor associated therewith. This is a combination of elements and circuit arrangement which greatly simplifies the circuitry and reduces the number of components necessary to provide a rhythm sound-producing device of the character described herein.

The drive circuit 227 includes an input terminal T ₃₀ connected to the base electrode of one of the transistors of the drive circuit. The drive circuit 229 includes a pair of input terminals T ₃₁ and T ₃₂ while the drive circuit 232 includes a pair of input terminals T ₃₃ and T ₃₄ . The drive circuit 236 includes an input terminal T ₃₅ while the drive circuit 238 includes an input terminal T ₃₆ . Similarly, the drive circuit 272 includes an input terminal T ₃₇ while the drive circuit 271 includes a plurality of input terminals T ₃₈ , T ₃₉ , and T ₄₀ . Certain ones of the input terminals T ₃₀ --T ₄₀ are direct current coupled to corresponding ones of output terminals of the time-point generator through the selector switch 26 and the matrix assembly 23, of FIG. 1. According to this invention, each of the transistors in the output circuits of the time-point generator are operated in the saturated mode of operation while each of the transistors in the input circuit of the instrument sound generator are operated in the cutoff mode of operation. This feature enables direct current coupling between the time-point generator and the instrument sound generator and greatly simplifies the circuitry as well as reducing the number of components necessary for proper operation thereof.

Also according to the present invention the multivibrator stage 226 includes a potentiometer 273 connected to the DC supply line thereby providing means for changing the frequency of oscillation of the circuit. By way of example, the multivibrator stage 226 is operated at approximately 70 Hz which is used to simulate the bass drum sound. However, as the sound of the bass drum may be changed somewhat by adjusting the tension of the skin of the drum so also can the sound of the bass drum generator 226 be adjusted.

The operation of each of the multivibrator circuits 220--226 together with their associated drive circuits are substantially the same and therefore a typical operation of one of the circuits will be given. By way of example, and not by way of limitation, the multivibrator circuit 226 produces a square wave output of 5 volts amplitude and a peak DC value of +5 volts. With no drive pulse supplied to input terminal T ₃₆ the transistor of the drive circuit 238 remains cut off and the capacitor connected in parallel with the transistor charges up to the peak signal value through diode 239 and the resistor 239a. This causes the junction between the diodes 239 and 259 to receive, for example, +5 volts thereby reverse biasing the diodes to the off condition so that no signal information from the multivibrator 226 passes therethrough. When a pulse from the time-point generator is applied to the terminal T ₃₆ through the matrices and selector switch, of FIG. 1, the transistor of drive circuit 238 is rendered heavily conductive and discharges the capacitor in parallel therewith thereby lowering the potential at the junction between diodes 239 and 259. While the point between the diodes 239 and 259 is below 5 volts, both diodes will conduct and signal information from the multivibrator 226 will pass therethrough. When the drive pulse from the time-point generator dissipates, the transistor of drive circuit 238 is again rendered nonconductive to cause the capacitor in parallel therewith to recharge to the peak signal voltage. This circuit action generates an envelope of signal information with an exponential decay controlled by resistor 239a and the capacitor in parallel with the transistor of the drive circuit 238.

The signal passing through the diodes 239 and 259 is filtered by the wave-shaping network 260 so as to cause the signal to approach a more perfect sinusoidal signal. The sinusoidal signal is then delivered to the base electrode of an amplifier transistor 253 which, in turn, delivers the audio signal information to the audio output circuit 262.

Circuit modifications are employed to obtain control of more than one multivibrator sound generator from a single drive circuit. For example, the multivibrators 223 and 224 are controlled simultaneously by the drive circuit 232. For purpose of simplicity, consider the drive circuit 232 a singular transistor type similar to that of the drive circuit 238. When an input pulse from the time-point generator is applied to the input terminal of the drive circuit the midpoints between the diodes 233 and 249 and diodes 234 and 251 decreases in voltage thereby causing the diodes to become conductive and simultaneously apply the output of the multivibrator 223 and 224 to the audio amplifier 243.

A still further circuit modification of the present invention is the use of two or more input circuits including a pair of timing capacitors separated by a diode so as to cause the diode keyers associated with certain ones of the sound generator multivibrators to be operative for different time intervals. For example, the drive circuit 232 includes a pair of transistors 232' and 232" which are connected to input terminals T ₃₃ and T ₃₄ respectively. Connected in parallel with the transistors is a capacitor 232a which, in turn, is connected in parallel with a capacitor 232b through a diode 232c. When a drive pulse is applied to either input terminal T ₃₃ or T ₃₄ the transistor associated therewith will be rendered conductive thereby discharging capacitor 232a and capacitor 232b through the diode 232c. That is, both capacitors 232a and 232b discharge simultaneously and at the same rate. However, the charging rate of the capacitors are different due to the blocking diode 232c. For example, the charging rate of capacitor 232a is controlled by the resistors connected to the midpoints between the gating diodes 233 and 249 and gating diodes 234 and 251. On the other hand, the charging rate of capacitor 232b has no effect on the charging rate of capacitor 232a due to the blocking diode 232c. It will be noted that the charging rate of capacitor 232a is made faster than the charging rate of capacitor 232b thereby maintaining a constant reverse bias on the diode 232c during the recharging operation to maintain the charge rates independent of one another.

The drive circuits 238 and 232, described hereinabove, are typical of the remaining drive circuits associated with the instrument sound generator of FIG. 3. It will be noted that the drive circuit 232 controls the multivibrators 223 and 224 simultaneously in response to the charge rate of capacitor 232a while the capacitor 232b controls the operation of one of the gating diodes associated with the noise generator 266.

Further circuit modifications are employed in the circuit arrangement of drive circuit 271. For example, the drive circuit 271 includes three input terminals T ₃₈ , T ₃₉ , and T ₄₀ . A drive pulse to the input terminal T ₃₉ discharges a capacitor 271' which recharges at a rate responsive to the value of resistance connected in series therewith. Similarly, an input pulse to terminal T ₃₉ causes forward bias of the gating diodes 270 but the forward conduction stops immediately upon removal of the drive pulse applied to terminal T ₃₉ as there is no capacitance to recharge in parallel with the transistor associated therewith. A drive pulse to terminal T ₄₀ discharges a pair of capacitors in parallel with the transistor associated therewith to again cause conduction through the gating diodes 270. Therefore, it will be understood that the drive circuit 241 controls a single gating circuit but at different time rates dependent on which terminal receives a drive pulse.

Seen in FIGS. 4 through 8 are typical matrices which are associated with the matrix assembly 23 of FIG. 1. It will be understood that the matrices shown in FIGS. 4--8 are shown to explain the circuit connection between the time-point generator and the instrument sound generator and are not to be construed in a limiting sense. The matrix may comprise a single resistor connected between one of the output terminals of the time-point generator and an input terminal of the instrument sound generator. However, a more sophisticated matrix is shown in FIG. 4 which includes a pair of terminals 400 and 401 for receiving drive pulses from the time-point generator. Also associated with the matrix of FIG. 4 is a plurality of terminals 402, 403, 404, and 405 for receiving gating pulses from the time-point generator. It will be understood that terminals represented by a circular designation indicate that the signal information applied thereto is a drive pulse while terminals designated by a square receive gating pulses. The matrix has a single output terminal 406 which is connected to one of the input terminals T ₃₀ -- T ₄₀ of FIG. 3 as desired. For example, a drive pulse applied to terminal 400 is delivered directly to the output terminal 406 through a resistor to render the drive circuit connected thereto operative. On the other hand, a drive pulse applied to terminal 401 passes through the first of a series of resistors and then through a diode 407 to ground potential as the circuit point 402 is connected to the collector of a saturated gating transistor, for example, terminal T ₂₀ of FIG. 2. Therefore, drive pulse applied to terminal 401 will not pass through the matrix until a positive gate pulse is applied to terminal 402. Gate pulses applied to terminals 403, 404, and 405 are used to drive the drive circuits through a diode 408. The diode 408 insures that gating signals applied to terminals 403--405 will not affect the drive pulse applied to terminal 400.

The matrix shown in FIG. 5 is a simple resistor matrix including a plurality of input terminals 500, 501, 502, 503, 504, and 505 connected to a single output terminal 506 through a plurality of corresponding resistors. A pulse or gate signal applied to any of the terminals of the matrix shown in FIG. 5 will cause the drive circuit connected to terminal 506 to be rendered operative.

Shown in FIG. 6 is a matrix including a plurality of terminals 600, 601, 602, and 603 arranged for connection to drive pulse output terminals of the time-point generator of FIG. 2. Also associated with the matrix of FIG. 6 is a pair of gating terminals 604 and 605 for connection to any one of the plurality of output gate terminals of the time-point generator. Terminals 600, 601, and 604 are resistive coupled to an output terminal 606 while terminals 602, 603, and 605 are resistive diodes coupled to the output terminal 606. Pulse signal information applied to terminals 600, 601, and 604 cause the drive circuit connected to output terminal 606 to be rendered operative. However, the pulse signal information applied to terminals 602 and 603 is shorted to ground potential through diode 607 until such time that a positive pulse is applied to terminal 605. The diode 608 is used to isolate the two portions of the matrix to eliminate interaction between terminals 600, 601, and 604 and terminals 602, 603, and 605.

FIG. 7 shows a further modified form of a matrix which may be used in accordance with the principles of this invention. The matrix of FIG. 7 includes a single pulse drive terminal 700 and a plurality of gate pulse terminals 701, 702, and 703. Pulse signal information will be applied to the output terminal 704 only when a positive pulse is applied to both terminals 702 and 703.

FIG. 8 shows a matrix including a pair of pulse input terminals 800 and 801 and a single gate terminal 802. The diode 803 is connected between a pair of resistors 804 and 805 which, in turn, are connected in parallel with a single resistor 806. The resistors 804 and 805 are selected to be approximately one-half the value of resistor 806 so that the total series resistance of the resistors is approximately equal to the value of resistor 806. A drive pulse applied to terminal 800 is delivered to the output terminal 807 to cause the drive circuit connected thereto to be rendered operative. Only a portion of the pulse signal applied to terminal 807 is passed to ground potential through the resistor 805 and diode 803. A drive pulse applied to terminal 801 is short circuited to ground potential when the transistor of the time-point generator connected to the terminal 802 is in the saturated mode. On the other hand, when a positive pulse is applied to the gate terminal 802 pulse signal information applied to terminal 801 will be delivered to output terminal 807.

According to the present invention a unique feature is the ability of the time-point generator to produce triplet pulses so as to obtain selectively different pulse width signal information so as to produce more realistic rhythm sounds. For a better understanding of this novel circuit arrangement reference is now made to FIG. 9 which shows the flip-flop circuits 59 and 60 in block form. Also shown in block form is the feedback amplifier 89, of FIG. 2. Input pulses to the flip-flop circuit 59 are applied thereto from the output of the flip-flop circuit 58, of FIG. 2, through a line 900. The feedback amplifier 89 is normally held in the saturated mode of operation by the bias applied thereto through the switch 19 which, in turn, is connected to a positive voltage source. Therefore, signals applied to the feedback amplifier 89 from the flip-flop circuit 60 through the line 901 have no effect on the output thereof, and the last two divider stages count in a normal divide by four fashion. When odd or divide-by-three function is required, the bias applied through switch 19 is removed by opening the switch. The removal of the bias now causes the feedback amplifier to be operated in an unsaturated mode to produce an inverted pulse in response to the output of the flip-flop circuit 60. That is, the output of the feedback amplifier follows the output of the last flip-flop circuit 60. Every third input cycle to the line 900 causes a pulse to be fed back from the flip-flop circuit 60 to the flip-flop 59 thereby resetting the flip-flop 59 and reducing the total count capacity of the two flip-flop circuits from four to three.

For a better understanding of the operation of the circuit arrangement of FIG. 9 reference is now made to FIG. 10 which shows the wave forms developed at various points throughout the circuit of FIG. 9. The wave form indicated by reference numeral 902 is representative of the pulse signal information applied to the input of the circuit through line 900. The wave form represented by reference numeral 903 indicates the wave form delivered from the output of flip-flop 59 to the input of flip-flop 60. The wave form represented by reference numeral 904 indicates the pulse signal information delivered from the feedback amplifier 89 to the input of flip-flop circuit 59 through the line 906. Additionally, the wave form represented by reference numerals 908 and 909 indicate the pulse signal information applied to output terminals 98 and 99 respectively. The broken line passing through the wave forms indicates the two modes of operation. That is, the wave forms to the left of the broken line, as seen on the drawing, indicate a divide-by-four condition while the wave forms on the right of the broken line indicate a divide-by-three condition. It will be noted that the output wave forms 908 and 909 are nonsymmetrical in a ratio of 2:1 during the divide-by-three mode of operation.

FIG. 11 is a detailed schematic diagram of an automatic follow circuit which may be used in conjunction with the rhythm sound-producing device of the present invention. The automatic follow circuit operates independently of the time-point generator and may be switched in and out of the circuit as desired and therefore may be considered optional equipment.

The automatic follow circuit shown in FIG. 11 includes an input terminal 300 which is connected to a primary winding 301 of a transformer 302. Connected in parallel with the primary winding 301 is a diode 303 in series with a resistor 304.

The secondary of transformer 302 includes a plurality of windings 306, 307, and 308. The winding 306 has the ends thereof connected to the base electrode of a pair of transistors 309 and 310, and a center tap connected to the emitter electrodes of transistors 309 and 310. Transistor 310 has the collector electrode thereof connected to one end of a secondary winding 311 of a sampled transformer 312. The transistor 309 has the collector electrode thereof connected to the base electrode of a transistor 313 and to one end of a capacitor 314. Transistor 313 is connected in cascade with a transistor 316.

The secondary winding 307 has the ends thereof connected to the base electrodes of a pair of transistors 317 and 318 and a center tap thereof connected to the emitter electrodes of the transistors. The collector electrode of transistor 317 is connected to one end of a primary winding 319 of the sampled transformer 312. The other end of the primary winding 319 is connected to a timing capacitor 320 and to the collector electrode of a transistor 321. The collector electrode of transistor 318 is connected to ground potential through a resistor 322 and to the secondary winding 308 through a resistor 323. The secondary winding 308 produces reset pulses which are used to control the timing of the automatic follow circuit.

The secondary winding 308 is connected to a plurality of diodes 326, 327, and 328. The diode 326 has the anode thereof connected to the base electrode of transistor 321. Also connected to the base electrode of transistor 321 is a capacitor 329 and one end of a resistor 330. The other end of resistor 330 is connected to a resistor 331. The emitter electrode of transistor 321 is connected to ground potential through a resistor 332. Resistors 330 and 331 are connected to a diode 333 and a resistor 334 connected in parallel therewith.

The collector electrode of transistor 321 is connected to the anode of a diode 336 and the collector electrode of a transistor 337. The cathode of diode 336 is connected to the cathode of diode 333 through a pair of capacitors 338 and 339. A resistor 340 is connected between the base and emitter electrodes of transistor 337, and a coupling capacitor 341 is connected between the base electrode of transistor 337 and a flip-flop circuit 342. It will be understood that the flip-flop circuit can be any one of the flip-flop circuit of the beat divider network of the time-point generator of FIG. 2. Preferably, the flip-flop circuit 342 corresponds to the flip-flop circuit 59 of FIG. 2a. The diode 328 has the anode thereof connected to one of the inputs of flip-flop 342 to reset the flip-flop in response to pulse signal information developed at secondary winding 308.

The diode 327 has the anode thereof connected to one of the base electrodes of a unijunction transistor 343 which forms part of a relaxation oscillator 344. Also connected to the oscillator 344 is the emitter electrode of transistor 316 through a resistor 346 and the base electrode of transistor 321 through resistors 330 and 347.

The output of flip-flop 342 is delivered to a flip-flop circuit 348, which corresponds to the flip-flop circuit 60 of the time-point generator of FIG. 2. The flip-flop circuit 348 is connected to the oscillator 344 through a switch 349, a diode 350 and a resistor 351. Pulse signal information developed by the oscillator 344 is applied to the first flip-flop 342 through the capacitor 352 and therefrom to the second flip-flop 348. The output of the flip-flop 348 is then delivered to a ring counter 353, corresponding to the ring counter shown in FIG. 2.

In operation, player-produced pulses are applied to the input terminal 300 to produce a current drive in the primary winding 301 of transformer 302. The transformer 302 will be referred to hereinafter as a sampling transformer. The current in the primary winding 301 produces a current pulse in the secondary windings 306, 307, and 308. The pulse in the secondary winding 308 applies a negative potential to the cathodes of diodes 326,327, and 328 while the pulse in secondary windings 306 and 307 cause the transistors associated therewith to become substantially short circuited. This action charges or discharges the timing capacitor 320 to a fixed potential point through the primary winding 319 of the sampled transformer 312 as well as completes the circuit of the secondary of the sampled transformer 312 between the input and output of transistors 313 and 316 which form an emitter follower stage. This action increases or decreases the charge on capacitor 314, depending on the state of charge existing on the capacitor, by an amount proportional to the difference in voltage on the timing capacitor 320 from the fixed potential. The negative pulse applied to the cathodes of diodes 326,327 and 328 momentarily cuts off the discharging transistor 321 and resets the unijunction oscillator 344 and the flip-flop 342.

One of the outputs of flip-flop circuit 348 is connected through the switch 349, diode 350 and the resistor 351 to the charging resistor of the unijunction oscillator 344 and to the output of the emitter follower transistor 316 through the resistor 346. The signal information applied to the oscillator 344 by the emitter follower 316 and by the feedback circuit of flip-flop 348 coact with one another to control the frequency of operation of the oscillator. With the switch 349 closed, conduction through the diode 350 and resistor 351, when flip-flop 348 is in one state lowers the voltage applied to the discharge transistor 321 and the unijunction oscillator 344. In the other state of operation of flip-flop 348 diode 350 is reversed biased and there is no conduction through the diode 350 and resistor 351. Therefore, the frequency of the unijunction oscillator 344 will be different for the two different conditions of flip-flop circuit 348. Therefore, this action causes a difference in length of the two halves of the square wave signal generated by flip-flop circuit 348 thereby providing the type of drive signal necessary for triplet timing in accordance with the present invention.

When the flip-flop circuit 342 is driven to the condition which does not trigger the input of flip-flop circuit 348, the charging transistor 337 is driven to saturation to fully charge the timing capacitor 320 to a positive potential. The capacitor 320 discharges at a rate depending upon the output of the emitter follower transistor 316 which controls the conduction of the discharge transistor 321. The capacitor 320 discharges until the voltage thereacross reaches a clamp point set by the other side of flip-flop circuit 342 through the clamping diode 336. The voltage on the timing capacitor 320 remains at the clamped potential until flip-flop circuit 342 changes to the other state of conduction which, in turn, lowers the voltage to the clamping diode 336. This change momentarily cuts off the discharge transistor 321 for a short period of time before it continues to discharge the timing capacitor 320.

For a better understanding of the discharge operation of timing capacitor 320 reference is now made to FIG. 13 which shows a wave form corresponding to the discharge range of the capacitor. The wave form 356 includes a first discharge portion 356a corresponding to the rate of discharge of capacitor 320 during the period of time when the clamping diode 336 is in the high state. The discharge rate terminates for a short period of time indicated by reference numeral 356b and then continues to discharge through transistor 321 as indicated by reference numeral 356c.

Seen in FIG. 12 is an alternate arrangement of an automatic follow circuit constructed in accordance with the principles of this invention. In FIG. 12, the transistor 318 has the collector electrode thereof connected to a capacitor 360 and to a resistor 361. Also connected to the resistor 361 is a second capacitor 362.

The collector electrode of transistor 318 and capacitor 360 are connected to one input of a midpoint detector circuit 363, while the capacitor 362 is connected to a second input of the midpoint detector 363. Also, the capacitor 362 is connected to the emitter electrode of a unijunction transistor 364 which includes a resistor 365 connected between one of the base electrodes and a positive voltage source. The midpoint detector 363 may take the form of a differential amplifier which is well known in the art.

In operation, the midpoint detector 363 produces pulses to drive the flip-flop 348. The midpoint detector 363 produces the pulses when the voltage across the unijunction transistor capacitor 362 becomes larger than the voltage across the capacitor 360. The capacitor 360 has a larger capacitance value than the capacitor 362 and connects thereto through the resistor 361 so that the voltage across capacitor 362 remains substantially at the midpoint voltage developed across capacitor 360. The primary winding 319 of the sampled transformer 312 is connected to the capacitors 360 and 362 through the transistors 317 and 318 when both transistors are in the conductive state. The circuit comprising switch 349, diode 350 and resistor 351 perform the same function as described in regard to FIG. 11.

The flip-flop circuit 348 triggers near the midpoint of the voltage swing on the unijunction capacitor 362 to mark the occurrence of the beat and AND beats. When a player-produced pulse is generated early for the beat or AND heat the voltage across capacitor 362 will be below the midpoint voltage and the sampled transformer secondary winding 311 will increase the charge on capacitor 314 thereby changing the output of the emitter follower transistor 316. On the other hand if the player-produced pulse is late, the voltage across the capacitor 362 is higher than the midpoint voltage and the charge across capacitor 314 is reduced. Therefore, the circuit arrangement provides means for sensing the difference between two voltages to produce a pulse in response thereto to control the operation of the time-point generator.

In accordance with the principles of this invention other modifications may be incorporated in the design of a clock generator which operates in accordance with the principles of this invention. Examples of such circuitry are shown in FIGS. 14 and 15.

The clock generator shown in FIG. 14 includes a unijunction transistor oscillator 11a. The oscillator 11a is formed by a unijunction transistor 410 which has the emitter electrode thereof connected to ground potential through a capacitor 411 and one of the base electrodes thereof connected to ground potential through a line 412. The other base electrode of transistor 410 is connected to a positive voltage source through a resistor 413. Also connected to the emitter electrode is a resistor 414 and a potentiometer 415. The potentiometer 415 is the tempo adjustment control for changing the frequency of oscillation.

The output of oscillator 11a is delivered through a capacitor 416 to a flip-flop circuit indicated generally by reference numeral 417. The flip-flop circuit 417 is of conventional circuitry including a pair of transistors 418 and 419. The emitters of transistors 418 and 419 are connected together and to ground potential. The base electrode of transistor 418 is connected to the collector electrode of transistor 419 through a capacitor 420 and resistor 421 which are connected in parallel. Similarly, the base electrode of transistor 419 is connected to the collector electrode of transistor 418 through a capacitor 422 and a resistor 423 which are connected in parallel. The collector electrode of transistor 418 is also connected to a resistor 424, and the collector electrode of transistor 419 is connected to a resistor 425. The resistors 424 and 425 are connected together and to a resistor 426 which, in turn, is connected to a positive voltage source.

According to the present invention, output pulses from the flip-flop circuit 417 are delivered to the base electrode of a transistor 427 via a selector switch 428. The selector switch 428 corresponds to the triplet switch 19, seen in FIG. 1. When switch 428 is closed, and a positive signal is applied to the base electrode of transistor 427, a capacitor 429 is connected in parallel with capacitor 411 thereby increasing the total charge capacitance of the oscillator circuit. By changing the charge capacitance of the oscillator circuit, the frequency of oscillation will remain the same but the period for one-half cycle of oscillation will change. This features provides the nonsymmetrical pulses necessary to generate the desired time-point signals for producing various rhythmic sounds.

Still another alternate form of a clock generator is shown in FIG. 15. The clock generator of FIG. 15 is formed by a free-running multivibrator designated generally by reference numeral 430. The multivibrator includes a pair of transistors 431 and 432 which have their emitter electrodes connected together and to ground potential, and their collector electrodes connected to a positive voltage source through a pair of resistors 433 and 434 respectively.

The base electrode of transistor 431 is connected to a resistor 435 and to a potentiometer 436 which forms the tempo adjustment control. The potentiometer 436 is also connected to the base electrode of transistor 432 through a resistor 437. The base electrode of transistor 432 is connected to the collector electrode of transistor 431 through a capacitor 438, and the base electrode of transistor 431 is connected to the collector electrode of transistor 432 through a capacitor 439.

According to the present invention, a resistor 440 and a selector switch 441 are connected in parallel with resistor 437. The selector switch 441 corresponds to the triplet switch 19, of FIG. 1. When switch 441 is open, the multivibrator circuit 430 produces symmetrical pulses at the output thereof for driving the ring counter and other pulse amplifiers. However, when switch 441 is closed, the resistance path formed by resistors 437 and 440 is decreased thereby changing the period of one-half cycle of the square wave generated by the multivibrator. Therefore, when switch 441 is open the multivibrator will produce symmetrical pulses, and when switch 141 is closed, the multivibrator will produce nonsymmetrical pulses.

Accordingly, the circuit arrangements shown in FIGS. 14 and 15 may be used in place of the tempo oscillator 11 and beat divider 12 which form the clock generator of FIG. 1.

Although the preferred embodiments of the present invention have been disclosed herein in detail, it will be understood that variations and modifications may be effected without departing from the spirit and scope of the novel concepts of this invention.