DETAILED DESCRIPTION

[0035] In the following detailed description of the invention, reference is made to the accompanying drawings which form a part hereof, and in which is shown, by way of illustration, specific embodiments in which the invention may be practiced. In the drawings, like numerals describe substantially similar components throughout the several views. These embodiments are described in sufficient detail to enable those skilled in the art to practice the invention. Other embodiments may be utilized and structural, logical, and electrical changes may be made without departing from the scope of the present invention.

[0036] The terms wafer and substrate used in the following description include any structure having an exposed surface with which to form the integrated circuit (IC) structure of the invention. The term substrate is understood to include semiconductor wafers. The term substrate is also used to refer to semiconductor structures during processing, and may include other layers that have been fabricated thereupon. Both wafer and substrate include doped and undoped semiconductors, epitaxial semiconductor layers supported by a base semiconductor or insulator, as well as other semiconductor structures well known to one skilled in the art. The term conductor is understood to include semiconductors, and the term insulator is defined to include any material that is less electrically conductive than the materials referred to as conductors. The following detailed description is, therefore, not to be taken in a limiting sense, and the scope of the present invention is defined only by the appended claims, along with the full scope of equivalents to which such claims are entitled.

[0037] FIG. 2 illustrates one technique to minimize the interwire capacitance, Cint, by using an intermediate line at ground for shielding. This technique is disclosed in a co-pending application by a common inventor, Dr. Leonard Forbes, entitled “Novel Transmission Lines for CMOS Integrated Circuits,” Ser. No. 09/364,199. The same is incorporated herein by reference.

[0038] Also, as disclosed in issued U.S. Pat. No. 6,255,852 by Dr. Leonard Forbes, entitled “Current Mode Interconnects on CMOS Integrated Circuits,” low impedance transmission lines such as those which exist on CMOS integrated circuits are more amenable to signal current interconnections over longer interconnection lines. U.S. Pat. No. 6,255,852 is incorporated herein by reference. These longer interconnection lines may be on the CMOS integrated circuit itself, an interconnection line between integrated circuits mounted in a module as for instance a memory module, an interposer upon which these integrated circuits are mounted, or on a printed circuit board upon which the integrated circuits are mounted. If the line is terminated with a low input impedance current sense amplifier then the line can be regarded as a transmission line terminated with the characteristic impedance of the interconnection line. This is advantageous in that:

[0039] (i) the signal delay depends only on the velocity of light on the line and is easily predictable and reproducible, eliminating or allowing for compensation for signal and/or clock skew,

[0040] (ii) there are no reflections at the receiving end of the line and this minimizes ringing,

[0041] (iii) noise signals will be smaller due to weaker coupling between lines resulting in better signal to noise ratios, the noise current will only be a small fraction of the signal current. The transmission of current signals rather than voltage signals is more desirable at high speeds, and in high speed or high clock rate circuits over longer interconnection lines. A CMOS circuit might for instance use a combination of techniques, conventional voltage signals over short interconnections with little coupling between lines and current signals over longer interconnections and where lines might be in close proximity.

[0042] FIG. 3A illustrates capacitive coupling between low impedance terminated interconnection lines. FIG. 3A illustrates signal transmission using correctly terminated transmission lines and current sense amplifiers, such as those disclosed in issued U.S. Pat. No. 6,255,852 by Dr. Leonard Forbes, entitled “Current Mode Interconnects on CMOS Integrated Circuits.” The signal interconnection or transmission line is terminated by the matching impedance of the current sense amplifier. This means the impedance looking into the sending end of the transmission line will just be the characteristic impedance of the line and the signal delay down the line will just be the small propagation delay down the line. The response time of the source follower being used as a line driver will be determined primarily by the longer rise time of the input voltage. This driver will supply a signal current whose rise time is basically just that of the input voltage signal.

[0043] FIG. 3A also illustrates the coupling to another signal line in close proximity, in this case the coupling will be both magnetic through the induced magnetic fields and mutual inductance and capacitive coupling. The noise current induced will be shown to be only a fraction of the signal current or the signal to noise ratio is high. Once received this signal current is converted back to a signal voltage by the current sense amplifier at the receiving end of the line. Since the signal propagation time is small, the signal delay time will in practice be limited by the rise time of the signal to the gate of the source follower. Since the gate capacitance of the source follower is small this can be very fast.

[0044] Other methods to minimize capacitive coupling between lines use low dielectric constant materials or insulators, or ground shields, such as shown in FIG. 2 . In the present invention, it is desirable to use very low impedance lines, it is also desirable to keep the capacitive coupling between lines small and the magnitude of voltage steps on the interconnection lines small. The current step will induce a voltage step at the load which is the magnitude of the load impedance times this current step. This voltage step while small, 1 mA times Zin in this example, still can induce a capacitively coupled noise signal on an adjacent line.

[0045] FIG. 3A shows an integrated circuit 300 in which a first transmission line, strip line, or coaxial line 301 A interconnects circuit components, e.g. a driver 310 to a receiver 320 . FIG. 3A illustrates a first transmission line 301 A over a conductive substrate 305 . Conventionally, a voltage signal (i.e. a 5 volt signal swing) is provided by the driver 310 to the transmission line 301 A. The schematic illustrations in FIG. 3A demonstrate that the transmission line 301 A includes a small resistance, shown generally by resistor symbols 302 A, 302 B, . . . , 302 N. Also, the transmission line 301 A includes a distributed inductance (L) which is represented generally by inductor symbols 303 A, 303 B, . . . , 303 N. In one embodiment, the driver 310 may be an inverter 310 and the receiver 320 may be an amplifier 320 . Capacitor plate symbols 304 (C) are used to schematically represent the capacitive coupling which occurs between the transmission line 301 A and the conducting substrate 305 . In FIG. 3A, a second transmission line 301 B is shown. Capacitor plate symbols 306 are used to schematically represent the capacitive coupling (Cint) which similarly occurs between the first transmission line 301 A and neighboring transmission lines, e.g. second transmission line 301 B.

[0046] FIG. 3B illustrates two interconnection lines in close proximity and the interwire capacitance between these lines and the mutual inductance coupling between the lines. (See generally, H. Johnson, “High-Speed Digital Circuits: A Handbook of Black Magic,” Prentice-Hall, 1993; and S. Ramo, J. R. Whinnery and T. Van Duzer, “Fields and Waves in Communication Electronics, 3rd Ed.,” John Wiley, New York, 1994). Although the interconnection lines on integrated circuits might tend to be more square than round, the concepts involved can be most conveniently described and formulas approximated by assuming for simplicity that the lines are round or circular. Approximate formulas have been developed describing round wires over conductive planes or two wires in close proximity, in this case they are interconnection wires on a CMOS integrated circuit, interposer, or printed circuit board.

[0047] In FIG. 3B the illustrated pair of interconnect, or transmission lines, 301 A and 301 B, displayed in a perspective view, are separated from a conducting substrate 305 . The transmission lines, 301 A and 301 B are spaced a distance (h) from the conducting substrate 305 and a distance (s) from one another. The transmission lines, 301 A and 301 B, are shown in a circular geometry, each with a diameter (a). Some general characterizations can be made about the transmission lines, 301 A and 301 B, in an environment floating or suspended in air. First, each transmission line, 301 A and 301 B, will have a characteristic impedance in air (Z ₀ ) approximately or generally given by Z ₀ ≅60 ln (4h/a). Second, each transmission line, 301 A and 301 B, has a inductance (L) which is L≅5.08×10 ⁻⁹ ×ln (4h/a) Henrys/inch (H/inch). Additionally, the two transmission lines, 301 A and 301 B, will exhibit an interwire mutual inductance (M) which is given by M=L×{1/[1+(s/h) ² ]}. Third, an interwire capacitive coupling (Cint) exists between the two transmission lines, 301 A and 301 B, and is expressed as Cint=πε/cos h ⁻¹ (s/a). Using the trigonometric relationship of cos h ⁻¹ (y)≅ln(2y), the interwire capacitive coupling can similarly be expressed as Cint≅πε/ln (2s/a). Thus, in this environment, the two transmission lines, 301 A and 301 B, exhibit an interline capacitance (Cint) given by Cint={0.7/[ln (2 s/a)]} pico Farads/inch (pF/inch). Lastly, each transmission line, 301 A and 301 B, will further exhibit capacitive coupling C with the conducting substrate 305 .

[0048] Again, in FIG. 3B the transmission lines, 301 A and 301 B, are spaced a distance (h) from the conducting substrate 305 . Using the method of images and the interwire capacitive relationship, Cint≅πε/ln (2 s/a), a single transmission line, 301 A, over a conducting substrate is given by C=2πε/ln (4h/a) pF/inch where h=s/2. Thus, in this environment, the two transmission lines, 301 A and 301 B, exhibit a capacitance, or capacitive coupling C with the conductive substrate 305 which is C≅{1.41/[In (4h/a)]} pF/inch. The above equations have been presented by assuming that the transmission lines have round or circular geometries. Actual transmission lines on integrated circuits might tend to be more square or rectangular than round due to present lithography techniques. Nevertheless, due to the actual physical size of transmission lines, determined according to minimum lithographic feature techniques, the formulas scale well to square, rectangular or other physical cross sectional geometries for the transmission lines.

[0049] The signal rise time (trise) in conventional voltage signaling is normally slow due to the limited ability of the transmission line drivers to supply the large currents needed to charge these capacitances over large voltage swings. The signal rise times are usually much larger than the signal transmission time down the line (tprop). Additionally, if two transmission lines are in close proximity then the voltage swing on one transmission line can induce a large voltage swing or noise voltage on the adjacent transmission line. The noise voltage is determined by the capacitance ratios of interwire capacitance, Cint, to the capacitance of the transmission line with the substrate, C. In other words, the noise voltage is determined according to the ratio Cint/C.

[0050] The values of Cint and C can be comparable, dependant upon the insulator thickness (h) under the transmission lines and the spacing between the transmission lines. Emphasis in prior art is placed upon minimizing the capacitance of the transmission line, C, by using thick insulators and low dielectric constant materials. Emphasis is also to some extent placed upon minimizing the interwire capacitance, Cint. Thus, the approach in the prior art results in a noise voltage which can be a large fraction of the signal voltage if the transmission lines are in close proximity and far removed from the substrate by being over thick insulators.

[0051] FIG. 4 is a perspective view illustrating a pair of neighboring transmission lines, 401 A and 401 B, above a conductive substrate 405 according to the teachings of the present invention. The present invention is designed to use current signaling across low impedance transmission lines, 401 A and 401 B, to reduce signal transmission delay and to improve signaling performance over longer transmission lines. Under conventional voltage signaling the current provided in the transmission lines is too weak to provide clean, accurately detectable current signal. In order to obtain better current signals in the transmission lines the signal to noise ratio of the transmission lines must be improved.

[0052] To improve the signal to noise ratio of the transmission lines, 401 A and 401 B, the capacitance coupling between the transmission lines, 401 A and 401 B, and the conductive substrate 405 , is made large. The characteristic impedance (Zo) of the transmission lines, 401 A and 401 B, can be expressed as Z ₀ ={square root}{fraction (L/C)}. Thus, making C large makes the characteristic impedance Zo=Zin, small and similarly makes the voltage division ratio for capacitive coupling small. In the present invention, C increases as the insulator 407 thickness (h) separating the transmission lines, 401 A and 401 B, from the ground plane, or substrate 405 is decreased. In FIG. 4 , the transmission lines, 401 A and 401 B, are separated a distance (h) from the conducting substrate 405 by an insulating layer 407 . In one embodiment, the insulating layer 407 is an oxide layer 407 . The capacitive coupling C between the transmission lines, 401 A and 401 B, and the conducting substrate 405 separated by an oxide layer 407 is given as C≅1.66/[ln(4h/a)] pF/cm. Additionally, the inductance (L) for the transmission lines, 401 A and 401 B, over the oxide layer 407 is L≅2×ln(4h/a) nanoHenrys/centimeter (nH/cm). The transmission lines, 401 A and 401 B, are shown in a square geometry having a width (a). The insulator 407 has a thickness (b) separating the transmission lines, 401 A and 401 B from the substrate. 405 . According to one embodiment of the present invention, the insulator thickness (b) is made thinner than the thickness (t) of the transmission lines, 401 A and 401 B. The center of the transmission lines, 401 A and 401 B, are a distance (h) above the conducting substrate 405 .

[0053] According to the teachings of the present invention, in one embodiment the thickness (b) of the insulator is equal to or less than 1.0 micrometers (μm). In one embodiment, the thickness (t) of the of the transmission lines, 401 A and 401 B is approximately equal to 1.0 micrometers (μm). In one embodiment, the thickness (t) of the transmission lines, 401 A and 401 B is less than 1.0 (μm). In one embodiment, the width (a) of the transmission lines, 401 A and 401 B is approximately 1.0 micrometers (μm). As one of ordinary skill in the art will appreciate upon reading the present disclosure, one embodiment of the present invention includes transmission lines 401 A and 401 B formed according to the above described dimensions and separated from the substrate 405 by an insulator having a thickness (b) of less than 1.0 micrometers (μm). In one exemplary embodiment, the transmission lines 401 A and 401 B have an input impedance (Z ₀ ) approximately equal to 50 ohms.

[0054] A co-pending application, by the same inventors, entitled “Capacitive Techniques to Reduce Noise in High Speed Interconnections,” application Ser. No. ______, describes minimizing interwire coupling capacitance, and making the insulator thickness over the group plane small, minimizing Zo. The same is incorporated herein by reference. According to the teachings described therein, a characteristic impedance of 50 ohms is easily realizable.

[0055] FIG. 5 is a schematic diagram for an interconnection on an integrated circuit 500 according to the teachings of the present invention. The interconnection on the integrated circuit 500 includes a pair of transmission lines, 501 A and 501 B, in close proximity. The first transmission line 501 A is separated by a distance (s) from the second transmission line 501 B. The first transmission line 501 A and the second transmission line 501 B each have a first end, 505 A and 505 B respectively. In one embodiment, the first end 505 A for the first transmission line 501 A is coupled to a driver 503 . The first transmission line 501 A and the second transmission line 501 B each have a second end, 506 A and 506 B respectively. In one embodiment, the second end 506 A is coupled to a termination 504 formed using a complementary metal oxide semiconductor (CMOS) process.

[0056] Reference to FIG. 5 is useful in explaining the reduced amount of noise current between two transmission lines, 501 A and 501 B, using the current signaling technique of the present invention. In one embodiment of the present invention, transmission lines, 501 A and 501 B, have a low characteristic impedances Zo. In one embodiment, the input impedance (Zin) seen by the driver 503 coupling to the first transmission line 501 A (in this example the “driven line”) is just the characteristic impedance Zo for the first transmission line 501 A. In other words, the CMOS termination 504 is impedance matched to the characteristic impedance Zo of the transmission line 501 A.

[0057] In one embodiment, the first transmission line 501 A is separated by approximately 3 μm from the second transmission line 501 B and the transmission lines have a length (l) of at least 500 μm. In another embodiment the transmission lines, 501 A and 501 B, have a length (l) of at least 0.1 cm, or 1000 μm. As in FIGS. 4 and 5 , the transmission lines, 501 A and 501 B, are separated from a conducting substrate by an insulating layer. In one embodiment, the insulating layer is an oxide layer. In this embodiment, the capacitive coupling C between the transmission lines, 501 A and 501 B, and the conducting substrate is given as C 1.66/[ln(4h/a)] pF/cm. In one exemplary embodiment, each transmission line, 501 A and 501 B, has a length (l) of 0.1 cm or 1000 μm, each has a width (a) of approximately 1.0 μm, and the insulator layer thickness (b) is approximately 0.2 μm. In this embodiment, the ln(4h/a) will be approximately 1. Thus, C≅1.66/[ln(4h/a)] pF/cm and for a line 0.1 cm long will produce a C≅0.2 pF. In the same embodiment, the inductance (L) for the transmission lines, 501 A and 501 B, over the oxide layer is L≅2×ln(4h/a) nH/cm, or L=0.2 nH for a line 0.1 cm long. In this embodiment, a 1 milli Ampere (mA) current step, i ₁ (t), is applied to the gate 502 of a transistor driver 503 . In one embodiment, the driver is an n-channel source follower driver 503 . In this embodiment, the rise time (trise) on the gate 502 of the driver 503 is approximately 100 ps. This is the limiting time on the system response since the signal delay (tprop) down a the transmission line is proportional to {square root}{fraction (L/C)}. For a 0.1 cm transmission line, 501 A or 501 B, tprop is only 7 ps. A current, di ₁ (t)/dt, of approximately 1×10 ⁷ A/sec is then produced on the first transmission line 501 A.

[0058] The noise current i ₂ (t) induced on the second transmission line 501 B by interwire capacitive coupling (Cint) is calculated as approximately i ₂ (t)=(Cint)×(V ₁ step/trise). The interwire capacitive coupling (Cint) between the transmission lines, 501 A and 501 B, separated by an oxide dielectric can be expressed as Cint=0.46 pF/cm. Again, for a 0.1 cm transmission line, 501 A or 501 B, Cint≅0.05 pF. As described in connection with FIG. 5, a 1 mA current provided to the first transmission line 501 A having a low characteristic impedance Zo of approximately 30 Ohms will result in a corresponding 30 mV Voltage step (V ₁ step) on the first transmission line 501 A. Therefore, if trise is 100 ps a noise current, i ₂ (t), of approximately 0.015 mA is produced on the second, neighboring, transmission line 501 B. This noise current, i ₂ (t), induced in the second transmission line 501 B is a very small percentage, or about 1%, of the signal current i ₁ (t) provided to the first transmission line 501 A. Hence, the signal to noise ratio (SNR) will be large. It can be shown, in general, that a signal to noise ratio (SNR) for the present invention, due to capacitive coupling is of the order (C/Cint) (trise/tprop); where, trise, is the rise time for the current signal and, tprop, the signal propagation time down the first transmission line 501 A. The rise time on the signal current, i ₁ (t), in the first transmission line 501 A is fast and just follows the rise time (trise) on the input signal, or 100 ps. The response time of this system utilizing current signals is thus much faster than those using voltage signals.

[0059] Reference to FIG. 5 is similarly useful to illustrate the noise voltage signal from magnetic coupling induced in the second transmission line 501 B by the signal current in the first transmission line 501 A. As shown in FIG. 5, a voltage will be induced in the second transmission line 501 B which has a magnitude that depends on the trise, di ₁ (t)/dt, of the current i ₁ (t) in the driven transmission line 501 A, and the mutual inductance coupling (M) between neighboring transmission lines, e.g. 501 A and 501 B. Each transmission line, 501 A and 501 B, has an inductance (L). As stated above, L≅0.2 nH for a 0.1 cm transmission line, 501 A and 501 B. In one exemplary embodiment, the current i ₁ (t) in the first transmission line, 501 A (in this example the “driven line”) rises to 1 mA in 100 ps. A current, di ₁ (t)/dt, of approximately 1×10 ⁷ A/sec is then produced on the first transmission line 501 A. As presented above in connection with FIGS. 3A and 3B , the mutual inductance coupling (M) can be expressed as M=L×{1/[1+(s/h) ² ]}. In one exemplary embodiment, s is approximately equal to 3 μm, and h is approximately equal to 0.7 μm. In this embodiment, M will equate to approximately M=0.02 nano Henrys (nH).

[0060] Using the relationship that the induced voltage (Vind)=M×di ₁ (t)/dt, Vind is approximately equal to 0.2 mV. During this 100 ps time period the induced voltage traveling down the second transmission line 501 B just sees the characteristic impedance Zo of the second transmission line 501 B. In one embodiment Zo is approximately 30 Ohms, so here, the current induced i ₂ (t) in the second transmission line is i ₂ (t)=Vind/Zo or 0.007 mA. This low value current is only approximately one percent (1%) of the signal current i ₁ (t) on the first transmission line, 501 A. Hence, a large signal to noise ratio (SNR) results. In contrast, under the prior technology, if high impedance capacitive loads had been used on high characteristic impedance lines and conventional voltage signaling employed there is typically a large noise voltage between the neighboring transmission lines, 501 A and 501 B. In the prior technology, the large noise voltage can be about one half as big as signal voltages.

[0061] The second transmission line 501 B has an equivalently rapid time constant, (L/R) to that of the first transmission line 501 A. In the embodiment presented above, the time constant is approximately 7 pico seconds (ps). The noise current i ₂ (t) in the second transmission line 501 B will reach a steady state in that time constant. The noise current stays at this steady state value until the end of trise, in this embodiment 100 ps, at which point i ₁ (t) stops changing. After this, the noise current in the second line decays away very quickly. Again, when the input impedance seen by the driver 503 is matched to the characteristic impedance Zo of the first transmission line 501 A, the signal to noise ratio (SNR) due to inductive coupling between the first transmission line 501 A and the second, or neighboring, transmission line 501 B is of the order, (L/M) (trise/tprop). In other embodiments, the actual mutual inductance and self inductances may vary from these given values without departing from the scope of the invention.

[0062] Inductive effects which become important at high speeds include not only the self inductance of the interconnection lines, L, but also the mutual inductance between lines, M. As shown with respect to FIG. 5 , previously the signal-to-noise ratio due to inductive coupling between lines is of the order, (L/M)(trise/tprop). Any technique which will minimize the mutual inductance between lines will improve the signal-to-noise ratio on long interconnection lines in integrated circuits with high switching speeds.

[0063] The present invention, as described further below, provides structures and methods through which inductive coupling on high speed interconnects can be further reduced thus increasing the signal to noise ratio across the same.

[0064] According to the teachings of the present invention, inductive coupling can be minimized by:

[0065] (i) magnetic shields above and below the lines

[0066] (ii) magnetic shields between lines

[0067] These magnetic shields may be:

[0068] (i) good conductors with a thickness greater than the skin depth, the conventional approach, but one which may not be possible or practical with interconnection lines of sub-micron dimensions

[0069] (ii) shields with high permeability metals to minimize the mutual coupling or inductance between lines

[0070] One embodiment of the invention, as discussed further below in connection with FIG. 6 , is a structure where an interconnection line is located between a ground buss and a power supply buss (which for the AC signal is AC ground) and as such constitutes a low impedance transmission line interconnection. If the ground and power supply busses are thicker than the skin depth at the frequency of interest, the electric and magnetic fields will be shielded and confined to the area between these plates. As shown in the embodiment of FIG. 6, a layered high permeability shielding line is placed between interconnection lines to distort the magnetic fields and shield the lines.

[0071] Other possible configurations are shown in FIGS. 7 - 12 . These configurations highlight the fact that a single metal might not have all the suitable properties for a given or desired implementation by systems designed for low noise operation. For example, two materials might be necessary, one which has the desired magnetic properties to confine magnetic fields and one to confine the electric fields. Accordingly, FIGS. 7 - 12 illustrate various alternative embodiments of the present invention as can be best suited to a particular system designed for low noise operation. These embodiments make use of a sandwich layer of both a high permeability material, well suited for magnetic shielding, as well as a low resistive conductive material that is well suited for electrical shielding. By placing even a thin layer of the high permeability material, a considerable amount of the magnetic field can be contained.

[0072] FIG. 6 illustrates one embodiment for a pair of neighboring transmission lines, 601 A and 601 B, according to the teachings of the present invention. FIG. 6 illustrates one or more transmission lines, shown as 601 A and 601 B. The one or more transmission lines, 601 A and 601 B, are spaced between a pair of electrically conductive planes 604 and 605 . As one of ordinary skill in the art will understand upon reading this disclosure, in one embodiment at least one of the electrically conductive planes is formed on a substrate. As one of ordinary skill in the art will understand upon reading this disclosure, the substrate can include an insulator, a semiconductor material, silicon on insulator material, or other materials. The invention is not so limited.

[0073] As shown in FIG. 6 , the invention includes a number of high permeability metal lines, shown in this embodiment as 602 A and 602 B. According to the teachings of the present invention, the number of high permeability metal lines, 602 A and 602 B, are formed of permalloy and Ni ₄₅ Fe ₅₅ films. As shown in FIG. 6 , the number of high permeability metal lines, 602 A and 602 B are interspaced between the one or more transmission lines, 601 A and 601 B. In one embodiment of the present invention, the one or more transmission lines, 601 A and 601 B, and the number or high permeability metal lines, 602 A and 602 B, are spaced parallel to one another and are oriented lengthwise perpendicular to the plane of the page illustrated in FIG. 6 . In the invention, the one or more transmission lines, 601 A and 601 B, and the number or high permeability metal lines, 602 A and 602 B, are separated from one another and from the pair of electrically conductive planes 604 and 605 by an insulator material 606 . In one embodiment of the present invention, the insulator material 606 includes an oxide.

[0074] In one embodiment as shown in FIG. 6 , the pair of electrically conductive planes 604 and 605 include metal ground planes 604 and 605 . In the invention, the electrically conductive planes, 604 and 605 , can be independently coupled to a ground source and/or a power supply bus as the same will be known and understood by one of ordinary skill in the art. In the embodiment shown in FIG. 6 , at least one of the pair of electrically conductive planes, 604 and 605 , is formed to a thickness (t) which is greater than a skin depth (sd) penetrable by electrically induced magnetic field lines.

[0075] As one of ordinary skill in the art will understand upon reading this disclosure, an electrical signal transmitted across the one or more transmission lines, 601 A and 601 B will induce a magnetic field surrounding the one or more transmission lines, 601 A and 601 B. In the embodiment of FIG. 6 such a magnetic field is illustrated by magnetic field lines 611 . According to the teachings of the present invention, the number of high permeability metal lines, 602 A and 602 B, and the electrically conductive planes, 604 and 605 , provide magnetic shielding to reduce the amount of magnetically induced noise on neighboring transmission lines, e.g. 601 A and 601 B.

[0076] FIG. 7 illustrates another embodiment for a pair of neighboring transmission lines, 701 - 1 and 701 - 2 , according to the teachings of the present invention. FIG. 7 illustrates one or more transmission lines, shown as 701 - 1 and 701 - 2 . The one or more transmission lines, 701 - 1 and 701 - 2 , are spaced between a pair of electrically conductive planes 704 and 705 . As one of ordinary skill in the art will understand upon reading this disclosure, any number of transmission lines, 701 - 1 , . . . , 701 -N, can be spaced between the conductive planes 704 and 705 . As one of ordinary skill in the art will understand upon reading this disclosure, in one embodiment at least one of the electrically conductive planes is formed on a substrate. As one of ordinary skill in the art will understand upon reading this disclosure, the substrate can include an insulator, a semiconductor material, silicon on insulator material, or other materials. The invention is not so limited.

[0077] As shown in FIG. 7 , the invention includes a number of high permeability metal lines, shown in this embodiment as 702 - 1 and 702 - 2 . According to the teachings of the present invention, the number of high permeability metal lines, 702 I and 702 - 2 , are formed of permalloy and Ni ₄₅ Fe ₅₅ films. As shown in FIG. 7 , the number of high permeability metal lines, 702 - 1 and 702 - 2 are interspaced between the one or more transmission lines, 701 - 1 and 701 - 2 . In one embodiment of the present invention, the one or more transmission lines, 701 - 1 and 701 - 2 , and the number or high permeability metal lines, 702 - 1 and 702 - 2 , are spaced parallel to one another and are oriented lengthwise perpendicular to the plane of the page illustrated in FIG. 7 . As one of ordinary skill in the art will understand upon reading this disclosure, any number of transmission lines, 701 - 1 , . . . , 701 -N can be spaced between any number of number high permeability metal lines, 702 - 1 , . . . , 702 -N. That is, one or more high permeability metal lines, 702 - 1 , . . . , 702 -N will separate one or more transmission lines, 701 - 1 , . . . , 701 -N. In the invention, the one or more transmission lines, 701 - 1 and 701 - 2 , and the number or high permeability metal lines, 702 - 1 and 702 - 2 , are separated from one another and from the pair of electrically conductive planes 704 and 705 by an insulator material 706 . In one embodiment of the present invention, the insulator material 706 includes an oxide.

[0078] In one embodiment as shown in FIG. 7 , the pair of electrically conductive planes 704 and 705 each include two layers, 704 A, 704 B and 705 A and 705 B. In this embodiment, a first layer, 704 A and 705 A respectively, include metal ground planes. A second layer or surface layer, 704 B and 705 B respectively, is formed of the same high permeability material as the number of high permeability metal lines, 702 - 1 and 702 - 2 . That is, the second layer or surface layer, adjacent to the one or more transmission lines, 701 - 1 and 701 - 2 , and the number of high permeability metal lines 702 - 1 and 702 - 2 , are formed of permalloy and Ni ₄₅ Fe ₅₅ films. As one of ordinary skill in the art will understand upon reading the present disclosure, the electrically conductive planes, 704 and 705 , can be independently coupled to a ground source and/or a power supply bus.

[0079] As one of ordinary skill in the art will understand upon reading this disclosure, an electrical signal transmitted across the one or more transmission lines, 701 - 1 and 701 - 2 will induce a magnetic field surrounding the one or more transmission lines, 701 - 1 and 701 - 2 . In the embodiment of FIG. 7 such a magnetic field is illustrated by magnetic field lines 711 . According to the teachings of the present invention, the number of high permeability metal lines, 702 - 1 and 702 - 2 , and the electrically conductive planes, 704 and 705 , provide magnetic shielding to reduce the amount of magnetically induced noise on neighboring transmission lines, e.g. 701 - 1 and 701 - 2 .

[0080] As shown in FIG. 7 , the second layer or surface layer, adjacent to the one or more transmission lines, 701 - 1 and 701 - 2 , and the number of high permeability metal lines 702 - 1 and 702 - 2 , each formed of permalloy and Ni ₄₅ Fe ₅₅ films, serve to shield the one or more transmission lines, 701 - 1 and 701 - 2 , from such electrically induced magnetic fields. The magnetic field lines 711 shown in FIG. 7 , illustrates the magnetic shielding effect provided by the number of high permeability metal lines, 702 - 1 and 702 - 2 , and the second layer or surface layer 704 B and 705 B, from magnetic fields produces by a current transmitted in the one or more transmission lines, 701 - 1 and 701 - 2 . As one of ordinary skill in the art will understand upon reading this disclosure, the first layer, 704 A and 705 A respectively, of the electrically conductive planes, 704 and 705 , provide a lower resistance such that there is very little resistance to the path of the return current.

[0081] As shown in the embodiment of FIG. 6 and other embodiments below, the second layer, or surface layer, 704 B and 705 B of high permeability metal, e.g. magnetic material permalloy and Ni ₄₅ Fe ₅₅ films, are formed on the inside of the conductive planes 704 and 705 , also referred to as the Vss or ground, adjacent to the one or more transmission lines, 701 - 1 and 701 - 2 . However as one of ordinary skill in the art will understand upon reading this disclosure, the second layer, or surface layer, 704 B and 705 B of high permeability metal can also be placed on the outside of the conductive planes 704 and 705 . As one of ordinary skill in the art will understand upon reading this disclosure, the number of high permeability metal lines 702 - 1 and 702 - 2 , each formed of permalloy and Ni ₄₅ Fe ₅₅ films, and the second layer, or surface layer, 704 B and 705 B of high permeability metal confine the magnetic fields in both the x and y direction. However, in this embodiment, the one or more transmission lines, 701 - 1 and 701 - 2 are only separated by a high permeability magnetic material that confines the magnetic field on both the x and y direction.

[0082] FIG. 8 illustrates another embodiment for a pair of neighboring transmission lines, 801 - 1 and 801 - 2 , according to the teachings of the present invention. FIG. 8 illustrates one or more integrated circuit lines, or transmission lines, shown as 801 - 1 and 801 - 2 . The one or more transmission lines, 801 - 1 and 801 - 2 , are spaced between a pair of electrically conductive planes 804 and 805 . As one of ordinary skill in the art will understand upon reading this disclosure, any number of transmission lines, 801 - 1 , . . . ., 801 -N, can be spaced between the conductive planes 804 and 805 . As one of ordinary skill in the art will understand upon reading this disclosure, in one embodiment at least one of the electrically conductive planes is formed on a substrate. As one of ordinary skill in the art will understand upon reading this disclosure, the substrate can include an insulator, a semiconductor material, silicon on insulator material, or other materials. The invention is not so limited.

[0083] As shown in FIG. 8 , the invention includes a number of electrically conductive metal lines, shown in this embodiment as 802 - 1 and 802 - 2 . According to the teachings of the present invention, the number of electrically conductive metal lines, 802 - 1 and 802 - 2 , include at least one surface layer 803 formed of a permalloy and Ni ₄₅ Fe ₅₅ film. As shown in the embodiment of FIG. 8 , the at least one surface layer 803 formed of a permalloy and Ni ₄₅ Fe ₅₅ film is formed on the number of electrically conductive metal lines, 802 - 1 and 802 - 2 , on opposing surfaces of the number of electrically conductive lines and adjacent to the number of integrated circuit lines, 801 - 1 and 801 - 2 . As shown in FIG. 8 , the number of electrically conductive metal lines, 802 - 1 and 802 - 2 , having at least one surface layer 803 formed of a permalloy and Ni ₄₅ Fe ₅₅ film, are interspaced between the one or more transmission lines, 801 - 1 and 801 - 2 . In one embodiment of the present invention, the one or more transmission lines, 801 - 1 and 801 - 2 , and the number or electrically conductive metal lines, 802 - 1 and 802 - 2 , are spaced parallel to one another and are oriented lengthwise perpendicular to the plane of the page illustrated in FIG. 8 . As one of ordinary skill in the art will understand upon reading this disclosure, any number of transmission lines, 801 - 1 , . . . , 801 -N can be spaced between any number of number electrically conductive metal lines, 802 - 1 , . . . , 802 -N, having at least one surface layer 803 formed of a permalloy and Ni ₄₅ Fe ₅₅ film. That is, one or more electrically conductive metal lines, 802 - 1 , . . . , 802 -N will separate one or more transmission lines, 801 - 1 , . . . , 801 -N. In the invention, the one or more transmission lines, 801 - 1 and 801 - 2 , and the number or electrically conductive metal lines, 802 - 1 and 802 - 2 , are separated from one another and from the pair of electrically conductive planes 804 and 805 by an insulator material 806 . In one embodiment of the present invention, the insulator material 806 includes an oxide.

[0084] In one embodiment as shown in FIG. 8 , the pair of electrically conductive planes 804 and 805 each include two layers, 804 A, 804 B and 805 A and 805 B. In this embodiment, a first layer, 804 A and 805 A respectively, include metal ground planes. A second layer or surface layer, 804 B and 805 B respectively, is formed of the same electrically conductive material as the at least one surface layer 803 on number of electrically conductive metal lines, 802 - 1 and 802 - 2 . That is, the second layer or surface layer, adjacent to the one or more transmission lines, 801 - 1 and 801 - 2 , and the number of electrically conductive metal lines 802 - 1 and 802 - 2 , are formed of permalloy and Ni ₄₅ Fe ₅₅ films. As one of ordinary skill in the art will understand upon reading the present disclosure, the electrically conductive planes, 804 and 805 , can be independently coupled to a ground source and/or a power supply bus.

[0085] As one of ordinary skill in the art will understand upon reading this disclosure, an electrical signal transmitted across the one or more transmission lines, 801 - 1 and 801 - 2 will induce a magnetic field surrounding the one or more transmission lines, 801 - 1 and 801 - 2 . In the embodiment of FIG. 8 such a magnetic field is illustrated by magnetic field lines 811 . According to the teachings of the present invention, the number of electrically conductive metal lines, 802 - 1 and 802 - 2 , having at least one surface layer 803 formed of a permalloy and Ni ₄₅ Fe ₅₅ film, and the electrically conductive planes, 804 and 805 , provide magnetic shielding to reduce the amount of magnetically induced noise on neighboring transmission lines, e.g. 801 - 1 and 801 - 2 .

[0086] As shown in FIG. 8 , the second layer or surface layer, adjacent to the one or more transmission lines, 801 - 1 and 801 - 2 , and the number of electrically conductive metal lines 802 - 1 and 802 - 2 , each formed of permalloy and Ni ₄₅ Fe ₅₅ films, serve to shield the one or more transmission lines, 801 - 1 and 801 - 2 , from such electrically induced magnetic fields. The magnetic field lines 811 shown in FIG. 8 , illustrates the magnetic shielding effect provided by the number of electrically conductive metal lines, 802 - 1 and 802 - 2 , having at least one surface layer 803 formed of a permalloy and Ni ₄₅ Fe ₅₅ film, and the second layer or surface layer 804 B and 805 B, from magnetic fields produces by a current transmitted in the one or more transmission lines, 801 - 1 and 801 - 2 . As one of ordinary skill in the art will understand upon reading this disclosure, the first layer, 804 A and 805 A respectively, of the electrically conductive planes, 804 and 805 , provide a lower resistance such that there is very little resistance to the path of the return current. As one of ordinary skill in the art will understand upon reading this disclosure, FIG. 8 shows a similar arrangement to that of FIG. 7 but both the electric and magnetic fields are now confined in both the x and y direction. Here the conductors are separated by not only a high permeability magnetic material but a sandwich of both a very low resistive ground plane which acts as a low resistive return path for induced currents (which is shown grounded) and high permeability magnetic material.

[0087] FIG. 9 illustrates another embodiment for neighboring transmission lines, 901 - 1 and 901 - 2 , according to the teachings of the present invention. FIG. 9 illustrates one or more integrated circuit lines, or transmission lines, shown as 901 - 1 and 901 - 2 . The one or more transmission lines, 901 - 1 and 901 - 2 , are spaced between a pair of electrically conductive planes 904 and 905 . As one of ordinary skill in the art will understand upon reading this disclosure, any number of transmission lines, 901 - 1 , . . . , 901 -N, can be spaced between the conductive planes 904 and 905 . As one of ordinary skill in the art will understand upon reading this disclosure, in